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International Journal of Microwave Science and Technology
Volume 2012 (2012), Article ID 157971, 7 pages
http://dx.doi.org/10.1155/2012/157971
Research Article

A Dual-Band SiGe HBT Frequency-Tunable and Phase-Shifting Differential Amplifier Employing Varactor-Loaded, Stacked LC Resonators

Graduate School of Electrical and Information Engineering, Shonan Institute of Technology, 1-1-25 Tsujido-Nishikaigan, Kanagawa, Fujisawa 251-8511, Japan

Received 7 July 2012; Accepted 24 September 2012

Academic Editor: Juan Carlos Bohórquez Reyes

Copyright © 2012 Kazuyoshi Sakamoto and Yasushi Itoh. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Abstract

A dual-band SiGe HBT frequency-tunable and phase-shifting differential amplifier has been developed for the future active phased array antennas with a multiband, multibeam, and multitarget tracking operation. The amplifier uses varactor-loaded, stacked LC resonators in the design of the output circuit in order to provide frequency-tunable and phase-shifting capabilities for dual frequencies. By utilizing the varactor-loaded LC resonator, which has a variable resonant frequency and a large insertion phase variation, frequency-tunable and phase-shifting performances become available. Moreover, by using the stacked configuration, the frequency and insertion phase can be varied independently for dual frequencies. A dual-band SiGe HBT differential amplifier has achieved a lower-frequency tuning range of 0.56 to 0.7 GHz for a higher fixed frequency of 0.97 GHz as well as a higher-frequency tuning range of 0.92 to 1.01 GHz for a lower fixed frequency of 0.63 GHz. A lower-frequency phase variation of 99° and a higher-frequency phase variation of 90.3° have been accomplished at 0.63 and 0.97 GHz, respectively. This is the first report on the dual-band differential amplifier with frequency-tunable and phase-shifting capabilities.

1. Introduction

Recent and future wireless systems require a wide range of data rates over several frequency bands, and thus adaptive and reconfigurable transceivers become necessary to support high performance and flexibility [1]. Meanwhile, microwave and millimeter-wave sensors and radar systems require multiband, multibeam scanning phased array antennas as well as a multi-target tracking capability [2]. To address these requirements, multifunction capabilities are strongly required. Especially multiband amplification, frequency tunability, gain and phase control are crucial for realizing adaptive and reconfigurable phased array systems. Most of the traditional multiband amplifiers, however, provide a single function [3]. Meanwhile, the authors have presented multiband amplifiers with frequency-tunable as well as gain control capabilities for bandpass and bandstop types [4, 5]. To enhance the operational capability, a novel design approach is proposed in this paper for the multiband low-noise differential amplifier with frequency-tunable and phase-shifting capabilities. In order to realize both frequency-tunable and phase-shifting performances, varactor-loaded LC resonators are used in the design of the output circuit of the differential amplifier. LC resonators are widely used in the reflection type phase shifter because they provide a full 360° phase shift [68]. With the use of the parallel LC resonator in the output shunt circuit, frequency-tunable and phase-shifting performances with high gain can be achieved at around resonant frequencies. In addition, by cascading the parallel LC resonators in a stacked form, frequency and insertion phases can be varied independently for multiple frequencies. As a multiband phase shifter, the resonator-switching or diplexer method has been proposed for the loaded-line type [9], reflection type [10], and LPF/HPF type [11]. These phase shifters, however, produce additional losses due to the switch or diplexer. Since the stacked LC resonator does not require additional switches or diplexers, a low insertion loss and a large phase variation can be expected. To make sure the usefulness of the novel design approach proposed in this paper, the dual-band SiGe HBT differential amplifier with frequency-tunable and phase-shifting capabilities has been actually designed, fabricated, and tested.

2. Circuit Design

A schematic diagram of the dual-band frequency-tunable and phase-shifting differential amplifier is shown in Figure 1. It employs varactor-loaded, stacked LC resonators in the design of the output circuit to achieve frequency-tunable and phase-shifting capabilities for dual frequencies. , , , and are inductors and capacitors consisting of LC resonators. and are a variable capacitance of the varactor diode which is controlled by and . is a decoupling capacitor having a large capacitance value. Resonant frequencies and are given as follows:

157971.fig.001
Figure 1: Schematic diagram of the dual-band frequency-tunable and phase-shifting differential amplifier.

When two parallel LC resonators are cascaded in a stacked form, the following bandstop frequency is generated. Consider

Now () then is necessarily sandwiched between and as

and are carefully designed in accordance with the target frequency range and phase-shifting amount. In order to realize and , several varactor diodes with a fixed capacitance ratio are connected in series or parallel.

A locus of the reflection coefficient of the varactor-loaded, stacked LC resonator is graphically shown in Figure 2. As the frequency increases from zero to , moves from the short to the open-circuited point . Since the LC resonator is actually lossy, the point has finite impedance. Then as the frequency increases up to , moves to the series resonant point where the impedance is not ideally short due to the lossy circuit. As the frequency increases from to , comes back to the point again. Finally goes to short. At around the resonant frequencies and , the insertion phase drastically varies from +180° to −180°. Therefore, with the precise control of and , the insertion phase can be tuned. In addition, at the resonant frequencies and , a peak gain of the amplifier can be obtained. The mechanism is depicted graphically in Figure 3 by using a frequency response of the amplifier.

157971.fig.002
Figure 2: Locus of the reflection coefficient of the varactor-loaded, stacked LC resonator.
157971.fig.003
Figure 3: Frequency response of the dual-band frequency-tunable and phase-shifting differential amplifier.

It is clearly demonstrated that the frequency can be varied with the coarse control of and and that the insertion phase can be also varied with the fine control of and . , , and are calculated by using from (1) to (3) for the circuit element values listed in Table 1. The circuit element values were chosen to meet the condition of (4). When varies from 6 to 15 pF, has a fixed value of 6 pF. Meanwhile, when varies from 6 to 15 pF, has a fixed value of 15 pF. The calculated results are shown in Figures 4 and 5, respectively. It is noted that and can be tuned independently and that is necessarily sandwiched between and.

tab1
Table 1: Circuit element values of Figure 1.
157971.fig.004
Figure 4: Calculated , , and ( varies from 6 to 15 pF. has a fixed value of 6 pF).
157971.fig.005
Figure 5: Calculated , , and ( varies from 6 to 15 pF. has a fixed value of 15 pF).

3. Circuit Fabrication

A photograph of the dual-band frequency-tunable and phase-shifting differential amplifier is shown in Figure 6.

157971.fig.006
Figure 6: Photograph of the dual-band frequency-tunable and phase-shifting differential amplifier: 16 × 16 × 1.2 mm3.

The differential amplifier was fabricated on the FR-4 substrate with a dielectric constant of 4.5. 0.35 m SiGe HBT with an of 25 GHz (Toshiba MT4S102T), Si varactor diode with a capacitance ratio of 2.5 : 1 (Toshiba 1SV279), 1005-type chip resistors, and inductors and capacitors are mounted on the substrate by soldering. The circuit size is 16 × 16 × 1.2 mm3.

4. Circuit Performance

4.1. Frequency-Tunable Performances

Measured gain, input and output return losses for a fixed and a variable are shown in Figures 7, 8, and 9, respectively. As a coarse tuning, was varied from 6 to 18 V with 1 V step since the anode voltage is equal to of 6 V. The maximal reverse voltage is 15 V. moves from 0.73 to 1.31 GHz for a fixed of 0.56 GHz. The gain varied from 0 to 7.2 dB for and keeps a constant value of 8.9 dB for . and were 6 V. Since the LC resonators are used in the output circuit, the output return loss is varied with the resonant frequency, but the input return loss keeps constant. Since the amplifier employed a reflective match approach for low-noise figure, the input and output return losses were less than 10 dB, which can be greatly improved by employing the lossy match approach using resistors [4]. In [4], the input and output return losses can be improved up to greater than 10 dB across the same frequency band by using the lossy match approach. This approach, however, provides a low gain and a poor noise figure. Thus, the reflective match approach was chosen. Meanwhile, the measured gain, input and output return losses for a variable and a fixed are shown in Figures 10, 11, and 12, respectively. In a similar way, was varied from 6 to 18 V with 1 V step. moves from 0.57 to 0.97 GHz and from 1.31 to 1.34 GHz. The gain varied from 8.7 to 13.1 dB for and 1.3 to 6.5 dB for . and were 6 V. If the varactor-loaded LC resonator shows an ideally open-circuited performance, and can be controlled independently. But actually the gain and frequency shift at was affected to a certain extent by controlling . This is because the parasitic resistance and low-Q factor of the varactor diode have made a serious effect on the gain and frequency at . Therefore, in addition to must be adjusted simultaneously for keeping constant. Moreover, as comes close to , the bandwidth at becomes wider, and the gain decreases in Figure 10. This is considered to be the same reason of the parasitic resistance and low-Q factor of the varactor diode. In Figures 7 and 10, the 3rd gain peaking can be seen around 1.6 GHz. This is mainly due to the fact that the LC resonator in Figure 1 shows a dual-band resonation due to the lead inductance of the varactor diode.

157971.fig.007
Figure 7: Measured gain for a fixed and a variable .
157971.fig.008
Figure 8: Measured input return loss for a fixed and a variable .
157971.fig.009
Figure 9: Measured output return loss for a fixed and a variable .
157971.fig.0010
Figure 10: Measured gain for a fixed and a variable .
157971.fig.0011
Figure 11: Measured input return loss for a fixed and a variable .
157971.fig.0012
Figure 12: Measured output return loss for a fixed and a variable .
4.2. Phase-Shifting Performances

Measured phase variations for a fixed and a variable are shown in Figure 13. As a fine tuning, was varied from 10 to 12 V with 0.2 V step. moves from 0.92 to 1.01 GHz for a fixed of 0.63 GHz. The phase variations at 0.97 GHz and 0.63 GHz were 90.3° and 2.5°, respectively. and were 6 V and 7 V, respectively. The gain variation, however, was 7 dB at 0.97 GHz. The gain variation can be improved by using a lossy match approach, which in turn degrades a phase-shifting amount; that is, there exists a design tradeoff between phase and gain variations. As one approach to address this problem, an additional variable gain amplifier (VGA) with a constant phase is utilized to compensate for the gain variation. Meanwhile, the measured phase variations for a fixed and a variable are shown in Figure 14. In a similar way, was varied from 6 to 8 V with 0.2 V step. moves from 0.56 to 0.7 GHz for a fixed of 0.97 GHz. The phase variation at 0.63 GHz was 99°. The phase variation at 0.97 GHz, however, was less than 2.1°. and were 6 V and 11 V, respectively.

157971.fig.0013
Figure 13: Measured phase variations for a fixed and a variable .
157971.fig.0014
Figure 14: Measured phase variations for a fixed and a variable .
4.3. Noise Figure Performances

Noise figure performances of the dual-band frequency-tunable and phase-shifting differential amplifier have been measured. The measured noise figures are plotted in Figure 15 for a fixed and a variable and in Figure 16 for a variable and a fixed . , , and were 6 V, 7 V and 9–13 V in Figure 15. Meanwhile, , , and were 6 V, 5–9 V, and 11 V in Figure 16. The measured noise figures were less than 4.1 dB at 0.63 GHz and less than 5.3 dB at 0.97 GHz in Figure 15. Meanwhile, the measured noise figures were better than 4.4 dB at 0.63 GHz and better than 4.3 dB at 0.97 GHz in Figure 16. As compared with Figures 7 and 10, a narrow variable range was chosen for and since it becomes difficult to show all noise figure data at the same time and make clear the difference. Overall the noise figure performances are not seriously affected by the frequency tuning.

157971.fig.0015
Figure 15: Measured noise figure performances for a fixed and a variable .
157971.fig.0016
Figure 16: Measured noise figure performances for a fixed and a variable .
4.4. IIP3

IIP3 performances were measured with two tones of a 10 MHz separation for a fixed (6 V) and a variable (6–18 V) as well as a variable (6–18 V) and a fixed (6 V). was 6 V. The amplifier has shown the minimum IIP3 of −8.7 dBm for a fixed (6 V) and a variable (6–18 V) as well as −6.5 dBm for a variable (6–18 V) and a fixed (6 V). A total collector current appeared in all measurements and ranged from 9 to 12 mA.

5. Conclusions

The dual-band SiGe HBT frequency-tunable and phase-shifting differential amplifier has been presented for use in the future active phased array antennas with a multiband, multibeam, and multi-target tracking operation. To realize multifunction capabilities including frequency-tunable and phase-shifting capabilities, the novel design approach to incorporate varactor-loaded, stacked LC resonators with different resonant frequencies into the circuit design has been proposed. Then to make sure the usefulness of the design approach, the dual-band frequency-tunable and phase-shifting differential amplifier using SiGe HBT and Si varactor diode have been actually designed, fabricated, and tested. It is confirmed from the measured results that the design approach proposed in this paper would be one of the candidates for the multifunctional amplifier for use in the current and future wireless, aerospace, microwave, and millimeter-wave sensors and radar systems.

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