A 25 kW, 25 kHz Induction Heating Power Supply for MOVPE System Using L-LC Resonant Inverter

Abstract

A topology named L-LC resonant inverter (RI) for induction heating (IH) applications takes most of the merits of the conventional series and parallel resonant schemes, while eliminating their limitations. In this paper, fundamental frequency AC analysis of L-LC RI is revisited, and a new operating point is suggested featuring enhanced current gain and near in-phase operation as compared to the conventional operating point. An approximate analysis of the circuit with square-wave voltage source is also described highlighting the effect of auxiliary inductor on the source current waveform. The analysis also leads to an optimum choice of the auxiliary inductance. The requirements of the metal organic vapour phase epitaxy (MOVPE) system in which a graphite susceptor is required to be heated to 1200°C demanding a 25 kW, 25 kHz IH power supply, the configuration of developed IH system, and experimental results are presented.

1. Introduction

The induction heating (IH) [1] is commonly used for heat treatment of metals (hardening, tempering, and annealing), heating prior to deformation (forging, swaging, upsetting, bending, and piercing), brazing and soldering, shrink fitting, coating, melting, crystal growing, cap sealing, sintering, carbon vapor deposition, epitaxial deposition, and plasma generation. IH is a noncontact method. The heat is generated only in the part, not in the surrounding area except by radiation. The location of the heating can be defined to a specified area on the metal component, thereby achieving accurate and consistent results. As heating occurs in the object itself, IH is considered more efficient than alternative methods.

An IH system comprises a basic induction power source which provides the required power output at the required power frequency, complete with matching components, an induction coil assembly, a method of material handling, and some method of cooling. Generally, full-bridge or half-bridge resonant inverters (RIs) are most commonly used as the power supplies for IH. The equivalent model of an IH coil with work piece can be represented in simplified form by an equivalent inductance (𝐿_eq) and resistance (𝑅_eq) as shown in Figure 1. If the IH coil is directly fed from a supply, the ratio of apparent to real power will be large. Therefore, the IH coil is properly compensated by capacitors and additional inductors in suitable configuration so that minimum reactive power is drawn from the source. Additionally to match the load voltage-current requirements to the available source, a matching network is required. The matching is normally achieved with the help of an isolation transformer of suitable turns ratio.

Figure 1 

Equivalent circuit representation of an IH coil.

Based on the connection of compensating capacitor with the IH coil, the two most commonly used RI topologies are as follows.(1)Series resonant inverter (SRI): the compensating capacitor is placed in series with the IH coil, and it is fed by a voltage source [2–6].(2)Parallel resonant inverter (PRI): the compensating capacitor is placed in parallel with the IH coil, and it is fed by the current source [7–11].

The analysis of these circuits has been performed in great details, and the comparative assessment is also reported in the literature [12, 13].

A topology named L-LC RI has been proposed for IH applications [14–21], which takes most of the merits of SRI and PRI while eliminating their limitations. It operates with input dc voltage source, thereby eliminating bulky input current smoothening inductor. It offers high current gain, which in turn reduces the current rating of the secondary winding of matching transformer and the feeder to coil.

Metal organic vapour phase epitaxy (MOVPE) [22] is a highly controlled method for the deposition of semiconductor epitaxial layers and heterostructures, which are required for the development of several optoelectronic and electronic devices. The process of MOVPE involves a vapour phase reaction between metalorganic compound and a hydride gas, which are transported to a heated (around 1200°C) graphite susceptor, resulting in a growth of the desired material. IH is one of the preferred methods for noncontact heating of the susceptor.

The paper aims to investigate the characteristics of L-LC RI for an application demanding a 25 kW, 25 kHz IH power supply to heat graphite susceptor to 1200°C in a MOVPE system for the growth of nitride semiconductors being developed at our Institute. Section 2 describes AC analysis of L-LC CN and investigates various characteristics when the converter is operated at the resonant frequency. The suggested operating point is different than the previously proposed operating point which provides enhanced current gain with smaller auxiliary inductance and results in near in-phase operation. Section 3 describes the operation of high-𝑄 L-LC compensating network (CN) with square-wave voltage source leading to an optimum choice of the auxiliary inductor. The requirements, description, and design of the practical system are discussed in Section 4. Experimental results are presented in Section 5.

2. Analysis of L-LC CN

Figure 2 shows the L-LC RI. Input dc source 𝑉_𝑑 could be an unregulated source (obtained with single-phase or three-phase diode rectifier and filter) or could be a regulated voltage source (obtained with single-phase or three-phase diode/SCR rectifier and filter or another front-end switch-mode regulator). In the former case, regulation of the power to the work piece should be done in the RI stage using frequency variation [23, 24], fixed-frequency pulse width modulation (PWM) [25–27], or pulse density modulation (PDM) [28, 29]. In the latter case, the output control can be done by varying 𝑉_𝑑, providing easy control of the output power over a wide range. However, two cascaded converters tend to reduce the overall efficiency.

Figure 2 

Circuit diagram of L-LC RI for IH application.

The following analysis based on the fundamental frequency approximation examines the important characteristics of L-LC topology. In an equivalent circuit of L-LC CN, shown in Figure 3, input voltage source is assumed to be a sinusoidal voltage source whose rms value is equal to the rms value of the fundamental component of the square-wave excitation. Following definitions are made for the analysis.

Figure 3 

Equivalent circuit diagram of L-LC RI for analysis.

Angular resonant frequency: 𝜔_𝑜=2𝜋𝑓_𝑜=1𝐿_eq𝐶. (1)

Normalized operating frequency: 𝜔_𝑛=𝜔𝜔_𝑜, (2) where 𝜔=2𝜋𝑓 is the angular operating frequency and 𝑓 is the operating or switching frequency.

Characteristic Impedance: 𝑍_𝑛=𝐿_eq𝐶. (3)

Circuit 𝑄: 𝜔𝑄=^𝑜𝐿_eq𝑅_eq=𝑍_𝑛𝑅_eq. (4)

Inductance ratio: 𝐿𝛾=^𝑎𝐿_eq. (5)

The expression for normalized current in coil and normalized source current (or the current in inductor 𝐿_𝑎) can be, respectively, derived as 𝐼_𝐿eq𝑁=𝐼_𝐿eq𝑉_𝑑/𝑍_𝑛=2√2𝜋1(1/𝑄)1−𝜔²_𝑛𝛾+𝑗𝜔_𝑛1+𝛾−𝜔³_𝑛𝛾,𝐼_𝐿𝑎𝑁=𝐼_𝐿𝑎𝑉_𝑑/𝑍_𝑛=2√2𝜋1−𝜔²_𝑛+𝑗𝜔_𝑛(1/𝑄)(1/𝑄)1−𝜔²_𝑛𝛾+𝑗𝜔_𝑛1+𝛾−𝜔³_𝑛𝛾. (6)

Next, the salient characteristics of the converter operating at the proposed operating point, 𝜔_𝑛=1, are investigated, leading to the following observations:

(1)𝐼_{𝐿eq𝑁𝜔𝑛=1}=2√2𝜋𝑄𝑄²+1−𝛾². (7) For 𝑄≫|(1−𝛾)|, 𝐼_{𝐿eq𝑁𝜔𝑛=1}≈2√2𝜋, (8)

(2)𝐼_{𝐿𝑎𝑁𝜔𝑛=1}=2√2𝜋1𝑄²+1−𝛾²,𝜙_𝜔𝑛=1=∠𝐼_{𝐿𝑎𝑁𝜔𝑛=1}=−tan⁻¹𝛾−1𝑄. (9)(10) The source current at 𝜔_𝑛=1 is also seen to be inductive for all values of 𝛾>1, and the phase angle is function of 𝑄 and 𝛾. If 𝛾=1, the source current is always in-phase with voltage.

Figure 4(a) shows the plot of 𝐼_𝐿eq𝑁 as a function of 𝛾 for different values of 𝑄. When the converter is operated at 𝜔_𝑛=1, 𝐼_𝐿eq𝑁 is seen to be relatively insensitive to 𝛾, particularly at high values of 𝑄—a typical operating condition in an IH application.

(a)

(b)

Figure 4 

The plots of 𝐼𝐿eq𝑁 (a) and 𝜙 (b) as a function of 𝛾 for different values of 𝑄 at two operating points.

Figure 4(b) shows the plot of 𝜙 as a function of 𝛾. The source current lags the applied square-wave voltage 𝜔_𝑛=1. While this lagging current is beneficial for zero-voltage-switching (ZVS) of the semiconductor switches, it can be anticipated that a higher value of 𝜙 results in higher source current causing more conduction loss in the switches. It is evident from the plots of Figure 4(b) that 𝜙 is small for operation at 𝜔_𝑛=1, particularly for operation with high 𝑄. Therefore, the source current and conduction loss in the switches will be smaller at this operating point.

The current gain of the CN, 𝐻, is given by 𝐼𝐻=^𝐿eq𝑁𝐼_𝐿𝑎𝑁=11−𝜔²_𝑛+𝑗𝜔_𝑛(),1/𝑄 (11) which can further be simplified at 𝜔_𝑛=1 as |𝐻|_𝜔𝑛=1=𝑄. (12) Figure 5 shows the plot of 𝐻 as a function of 𝜔_𝑛 for 𝛾=5 and different values of 𝑄. Maximum current gain (equal to 𝑄) is observed for operation at the proposed operating point 𝜔_𝑛=1.

Figure 5 

The plots of 𝐻 as a function of 𝜔𝑛 for 𝛾=5.

The characteristics and design of L-LC RI for IH applications operating at 𝜔_𝑛=𝜔_{𝑜_𝑐}=(1+𝛾)/𝛾 have been reported extensively in the literature [15–21]. At this operating point, |𝐻|_{𝜔𝑛=𝜔𝑜_𝑐}=𝑄𝛾𝑄².+𝛾1+𝛾 (13) For 𝑄≫𝛾, |𝐻|_{𝜔𝑛=𝜔𝑜_𝑐}≈𝛾, which is less than theoretical maximum value given by (12) at the proposed operating point. Thus, as also shown in Figure 5, the proposed operating point of L-LC RI results in the enhanced current gain. Further, the expression for 𝜙 at the conventional operating point can be written as 𝜙_{𝜔𝑛=𝜔𝑜_𝑐}=tan⁻¹−𝛾𝑄1+𝛾𝛾. (14) The plots of (14) as a function of 𝛾 are shown in Figure 4(b) for 𝑄=5 and 𝑄=20. It can be seen that phase angle is smaller at the proposed operating point than the conventional operating point, thereby resulting in less conduction loss in the switches.

3. Behaviour of L-LC CN with Square-Wave Voltage Source

Analysis based on the fundamental frequency approximation presented in Section 2 suggests that the converter, when operated at 𝜔_𝑛=1 offers highest current gain and small phase angle between source voltage and current resulting in low reactive power loading and low conduction loss in the semiconductor devices of the square-wave inverter. The analysis presented in Section 2 assumes sine wave voltage source at the input. In practice, the input voltage is a square- wave generated by operating switches 𝑆₁--𝑆₄ in Figure 2 at 50 per cent duty cycle. In this section, the behaviour of L-LC resonant inverter for operation at 𝜔_𝑛=1 with square-wave voltage source examined. Equivalent circuit for L-LC RI for analysis with square-wave voltage source is shown in Figure 6(a). The source voltage 𝑣_𝑠 can be defined as 𝑣_𝑠(𝑡)=𝑉_𝑑⋅signsin𝜔_𝑜𝑡, (15)𝑣_𝑠 can be decomposed into its fundamental component 𝑣_𝑠1 and harmonics 𝑣_ℎ as 𝑣_𝑠(𝑡)=𝑣_𝑠1(𝑡)+𝑣_ℎ(𝑡)=4𝑉_𝑑𝜋sin𝜔_𝑜𝑡+𝑣_ℎ(𝑡). (16) From (15) and (16) 𝑣_ℎ(𝑡)=𝑉_𝑑⋅signsin𝜔_𝑜𝑡−4𝑉_𝑑𝜋sin𝜔_𝑜𝑡=𝑉_𝑑−4𝑉_𝑑𝜋sin𝜔_𝑜𝑡𝑇for0≤𝑡≤^𝑠2=−𝑉_𝑑−4𝑉_𝑑𝜋sin𝜔_𝑜𝑡𝑇for^𝑠2≤𝑡≤𝑇_𝑠, (17) wherein 𝑇_𝑠 is the switching period. Thus, equivalent circuit of Figure 6(a) can be redrawn as Figure 6(b), wherein 𝑣_𝑠 decomposed into 𝑣_𝑠1 and 𝑣_ℎ.

(a)

(b)

(c)

(d)

Figure 6 

(a) Equivalent circuit for L-LC RI for analysis with 𝑣𝑠. (b) Equivalent circuit redrawn with 𝑣𝑠 decomposed into 𝑣𝑠1 and 𝑣ℎ. (c) Approximate equivalent circuit for fundamental frequency. (d) Approximate equivalent circuit for harmonics frequencies.

Equivalent impedance (𝑍_eq) of the 𝐿_eq, 𝑅_eq, and 𝐶 resonant circuit at 𝜔_𝑛=1 is given by 𝑍_eq𝜔𝑛=1=𝑅_eq𝑄1+𝑄²≈𝑅_eq𝑄²for𝑄≫1. (18) Further, for high-𝑄 resonant circuit in IH applications, 𝑍_eq can be approximated to be zero for all operating points other than 𝜔_𝑛=1. Under this assumption, equivalent circuit for Figure 6(b) can be decoupled for operation at 𝜔_𝑛=1 and at harmonic frequencies, as shown in Figures 6(c) and 6(d), respectively. Assuming 𝑅_eq𝑄²≫𝜔_𝑜𝐿_𝑎, equivalent circuit at 𝜔_𝑛=1 can further be simplified as shown in Figure 6(c). With this simplification, the circuit can be analyzed for fundamental and harmonic frequencies separately to find out individual source current and then be added to find out resultant source current. An expression for 𝑖_𝑠 can be derived as 𝑖_𝑠(𝑡)=𝑖_𝑠1(𝑡)+𝑖_ℎ(𝑡),𝑖_𝑠1(𝑡)=4𝑉_𝑑𝜋𝑄²𝑅_eqsin𝜔_𝑜𝑡,𝑖_ℎ(𝑡)=𝑉_𝑑𝐿_𝑎𝑇𝑡−^𝑠4−4𝑉_𝑑𝜋𝜔_𝑜𝐿_𝑎sin𝜔_𝑜𝜋𝑡−2𝑇for0≤𝑡≤^𝑠2𝑉=−^𝑑𝐿_𝑎𝑡−3𝑇_𝑠4−4𝑉_𝑑𝜋𝜔_𝑜𝐿_𝑎sin𝜔_𝑜𝜋𝑡−2𝑇for^𝑠2≤𝑡≤𝑇_𝑠. (19) Figure 7(a) shows the calculated waveforms for the operation of the circuit under low-𝑄 condition. It can be easily noticed that 𝑖_𝑠1(𝑡), being inversely proportional to 𝑄, is significantly larger than 𝑖_ℎ(𝑡) for low-𝑄 values. Therefore, resulting 𝑖_𝑠(𝑡) is also nearly sinusoidal. Under this condition, the predictions of the ac analysis (Section 2) are reasonably accurate. However, under high-𝑄 condition, as shown in Figure 7(b), amplitude of 𝑖_𝑠1(𝑡) is much smaller than 𝑖_ℎ(𝑡). Therefore, 𝑖_𝑠(𝑡) is nearly the same as 𝑖_ℎ(𝑡) and is highly nonsinusoidal. Under this condition, the predictions of Section 2 tend to be erroneous. Since the analysis of Section 2 does not account for harmonics, the actual peak and rms values of 𝑖_𝑠 calculated by (19) are significantly more than those predicted by (9). This will result in degradation of the current gain from its value predicted by (12). However, the amplitude and rms value of 𝑖_ℎ(𝑡) can be controlled by choosing a proper value of 𝐿_𝑎. Figure 7(c) shows the calculated waveform of 𝑖_𝑠(𝑡) for different values of 𝛾 when 𝑄=50. Since the rms value of 𝑖_ℎ(𝑡) can be reduced by increasing the value of 𝐿_𝑎, the degradation in the current gain can be corrected to some extent by choosing higher value of 𝛾. Figure 8 shows the plots of current gain as a function of 𝑄 for different values of 𝛾. A plot of the current gain predicted by (12) is also shown for direct comparison. The figure also shows the evolution of the current gain at the conventional operating point predicted by (13) for 𝛾=1 and 𝛾=5. It is noticed that the actual current gain at the proposed operating point is higher than that at the conventional operating point.

(a)

(b)

(c)

Figure 7 

Typical calculated waveforms of 𝑣𝑠(𝑡), 𝑖𝑠1(𝑡), 𝑖ℎ(𝑡), and 𝑖𝑠(𝑡) for low-𝑄 (𝑄=1;𝛾=1) (a) and high-𝑄 (𝑄=50;𝛾=1) (b) operation. The effect of 𝛾 on the waveform of 𝑖𝑠(𝑡) is illustrated by the calculated waveforms in (c) for 𝑄=50.

Figure 8 

The plots of current gain as a function of 𝑄 for different values of 𝛾 at the at the proposed operating point 𝜔𝑛=1 and conventional operating point 𝜔𝑛=𝜔𝑜_𝑐.

The physical size of an inductor depends on its value, peak current, and rms current. Since, peak and rms value of 𝑖_𝑠(𝑡) decreases with increasing 𝐿_𝑎, it is intuitive to expect that the physical size of 𝐿_𝑎 will first reduce as 𝐿_𝑎 increased, reach a minimum and subsequently increase with further increase in 𝐿_𝑎. To obtain a value of 𝐿_𝑎 resulting in minimum size, the following term is defined as its normalized size index: 𝐾_𝐿𝑎=𝐿_𝑎𝐼_𝑠,pk𝐼_𝑠,rms𝐿_eq𝐼_𝐿eq,pk𝐼_𝐿eq,rms. (20) Figure 9 shows the plots of (20) for different values of 𝑄. It is seen that for high-𝑄 applications, 𝐿_𝑎 should typically be 5 times 𝐿_eq.

Figure 9 

The plots of (20) showing the variation of 𝐾𝐿𝑎 as a function of 𝛾 for different values of 𝑄.

4. System Description

MOVPE is a highly controlled method for the deposition of semiconductor epitaxial layers and heterostructures, which are required for the development of several optoelectronic and electronic devices. The process of MOVPE involves a vapour phase reaction between metalorganic compound and a hydride gas, which are transported to a heated (around 1200°C) graphite susceptor, resulting in a growth of the desired material. A power supply is required to heat graphite susceptor (50 mm diameter and 20 mm length) to 1200°C in a quartz reactor of 80 mm diameter.

4.1. Work Coil and Resonant Capacitor

While recommended coil design practices suggest that the coil should be as close to the susceptor and the coil length should be larger than that of the susceptor [1], physical dimensions of the susceptor and the reactor force the inner diameter of the coil to be 100 mm. Further, since a stainless steel showerhead with nozzles for flowing gases into the reactor is attached to one end of the reactor, the maximum length of the coil is also limited to 50 mm. The coil is made up of hollow copper conductor. The number of turns, conductor diameter, and its wall thickness are optimized considering various parameters like coil resistance, power loss, electrical efficiency, required cooling water flow rate, and pressure drop. The designed coil has 4 turns of hollow copper conductor with 3/8 inch OD and SWG 19 wall thickness. The coil resistance is estimated to be 11.3 mΩ, and coil inductance is 2.70 μH. The operating frequency of 25 kHz has been fixed to ensure uniform heating of the susceptor (the skin depth of graphite is nearly 17 mm at 25 kHz). Equivalent resistance of work piece is estimated to be 16.6 mΩ. The coil is designed to carry a maximum of 1000 A rms.

A bank of 6 conduction-cooled capacitors of 3 μF each has been used as the resonant capacitor 𝐶. These capacitors are mounted on a cold plate, which in turn is water cooled. However, only 5 capacitors are used to get the resonant frequency of 25 kHz. The capacitor bank is kept very close to the coil to reduce the circulating path.

4.2. Power Circuit

A schematic circuit diagram of the power circuit is shown in Figure 10, which can broadly be divided into three sections: front-end-three phase diode rectifier with filter, a dc-dc converter, and the L-LC RI. In addition to these, the actual power circuit also consists of breakers, EMI filter, and inrush-current-limiting circuit in the rectifier stage. However, these are not shown in Figure 10 for clarity.

Figure 10 

The power circuit and the control block diagram of a 25 kHz, 25 kW induction heating power supply using L-LC RI for MOVPE system. #The process controller, shown for the completeness, is not used in the laboratory testing of the system.

The input to the power supply is 415 V, 50 Hz, and 3-phase ac. The values of filter inductor (𝐿_𝑓) and capacitor (𝐶_𝑓) in the input rectifier section are 2.2 mH and 3 mF, respectively, giving the cut-off frequency of 70 Hz.

It is seen in Section 2 that the L-LC RI exhibits desired behaviour only when it is operated at 𝜔_𝑛=1. Therefore, variation of switching frequency for output power control is not permitted. Therefore, fixed-frequency control methods such as pulse width modulation (PWM) or the quantized control methods such as pulse density modulation can be used. However, the soft-switching operation of the switches in RI stage may not be guaranteed over the entire operating range, and the quantized control methods, such as PDM, result in discrete output levels and also have limited range of output power control. If, on the other hand, input voltage to the RI stage is controlled, output power control over a wider range and operational flexibility can be ascertained. Demerits of this scheme, namely, requirement of an intermediate dc-dc converter stage and reduction of overall conversion efficiency, are sacrificed for aforementioned advantages.

A dc-dc buck converter is chosen as the intermediate dc-dc converter stage. IGBT 𝑆_𝑏 and diode 𝐷_𝑏 constitute the main switching cell. A passive loss-less snubber circuit (composed of snubber inductor 𝐿_𝑠, capacitors 𝐶_𝑠1, 𝐶_𝑠2, and diodes 𝐷_𝑠1, 𝐷_𝑠2, and 𝐷_𝑠3) is used to limit switching losses in this stage. It is worthwhile to note that apart from the careful choice of the component values, described in [30], an equal attention to several practical aspects [31], such as forward recovery of snubber diodes and stray inductances, is also important for effective snubber action. Inductor 𝐿_𝑏, capacitors 𝐶_𝑏, 𝐶_𝑏𝑑, and resistor 𝑅_𝑏𝑑 constitute a high-frequency damped low-pass filter.

Two half-bridge IGBT modules SKM100GB123D are used to realize the H-bridge in the inverter section feeding square- wave voltage to the high-frequency isolation transformer. The transformer 𝑇_𝑟 having 1 : 1 turns ratio has been developed using 7 pairs of EE80 core with 7 turns of the primary winding and 7 turns of the secondary winding. While the primary is wound using 0.2×50mm copper foil, oxygen-free high conductivity hollow copper conductor is used for water-cooled secondary winding. The results for Figure 9 suggest that the additional resonant inductor 𝐿_𝑎 should approximately be 5 times 𝐿_eq for high-𝑄 operation. Therefore, in the present system, 𝐿_𝑎=13.5𝜇H is chosen. The transformer leakage inductance is estimated to be 5 μH. The additional 8.5 μH inductance is realized by running the secondary turns around auxiliary gapped core (2 pairs of EE80), following the configuration suggested in [32].

A capacitor has been placed in series with the transformer primary (not shown in Figure 10) to prevent the transformer saturation in the event of unsymmetrical excitation. Table 1 summarizes the values of the components and part numbers of the semiconductor devices used in the system.

Table 1 

Component values and device part numbers used in the system.

4.3. Control Circuit

The input dc voltage to the square-wave inverter is controlled to regulate the susceptor temperature. The block diagram of overall control system is also shown in Figure 10. A thermocouple-type temperature sensor and process PID controller module are planned to be used for programming and controlling susceptor temperature. The output of PID module acts as the reference for the inner control loop to control the input dc voltage to the square-wave inverter by controlling the duty ratio of the buck converter. An additional phase control loop is employed to maintain the tune-in condition of the RI stage against the slow drift of resonant frequency over varied operating conditions and time. This loop senses the phase of the inverter output current using a phase detector, PD. The output of the controller drives a voltage controlled oscillator (VCO) that adjusts the switching frequency of the inverter in such a way that the inverter output current lags the voltage slightly.

5. Results

The photograph of the developed induction heating power supply, being tested in laboratory to heat a graphite susceptor, is shown in Figure 11. The graphite susceptor heated to 1200°C in air is shown in the inset.

Figure 11 

The photograph of the developed induction heating power supply. The graphite susceptor heated to 1200°C is shown in the inset.

The inverter output voltage, 𝑣_𝑠, (trace 1, 200 V/div) and current waveforms, 𝑖_𝑠, (trace 2, 25 A/div) are shown in Figure 12(a). The nature of current waveform matches with the predicted waveform given in Figure 7. The waveforms of collector-to-emitter (trace 1, 200 V/div) and gate-to-emitter (trace 2, 10 V/div) voltage of an IGBT in H-bridge inverter during turn-on and turn-off transitions are shown in Figures 12(b) and 12(c), respectively, demonstrating the soft-switching. Figure 12(d) shows the waveforms of 𝑣_𝑠 (trace 1, 200 V/div) and the voltage across the work coil (trace 2, 200 V/div) showing that only fundamental component of input square-wave voltage has been passed to the work coil.

(a)

(b)

(c)

(d)

Figure 12 

Experimental waveforms. (a) 𝑣𝑠 (trace 1, 200 V/div) and 𝑖𝑠 (trace 2, 25 A/div); 𝑥-scale: 5 μs/div. (b) Waveforms at turn-on: IGBT collector-to-emitter voltage (trace 1, 200 V/div) and gate-to-emitter voltage (trace 2, 10 V/div) 𝑥-scale: 500 ns/div. (c) Waveforms at turn-off: IGBT collector-to-emitter voltage (trace 1, 200 V/div) and gate-to-emitter voltage (trace 2, 10 V/div) 𝑥-scale: 500 ns/div. (d) 𝑣𝑠 (trace 1, 200 V/div) and the voltage across the work coil (trace 2, 200 V/div) 𝑥-scale: 10 μs/div.

6. Conclusion

The paper reports the various issues in the development of an IH power supply for MOVPE application using L-LC RI. The fundamental frequency ac analysis of L-LC RI is revisited, and it is shown that the converter exhibits enhanced current gain and near-unity power factor operation when operated at the resonant frequency. Further analysis of the circuit with square-wave voltage source highlights the effect of the auxiliary inductor on the source current waveform leading to the optimum choice for the value of auxiliary inductor. The requirements of MOVPE system demanding a 25 kW, 25 kHz IH power supply to heat a graphite susceptor to 1200°C, the configuration of developed IH system, and experimental results are presented, thereby demonstrating the suitability of L-LC RI operating at its resonant frequency for this application.