Dual Strip-Excited Dielectric Resonator Antenna with Parasitic Strips for Radiation Pattern Reconfigurability

Abstract

A novel pattern reconfigurable antenna concept utilizing rectangular dielectric resonator antenna (DRA) placed over dielectric substrate backed by a ground plane is presented. A dual strip excitation scheme is utilized and both excitation strips are connected together by means of a 50 Ω microstrip feed network placed over the substrate. The four vertical metallic parasitic strips are placed at corner of DRA each having a corresponding ground pad to provide a short/open circuit between the parasitic strip and antenna ground plane, through which a shift of in antenna radiation pattern in elevation plane is achieved. A fractional bandwidth of approximately 40% at center frequency of 1.6 GHz is achieved. The DRA peak realized gain in whole frequency band of operation is found to be above 4 dB. The antenna configuration along with simulation and measured results are presented.

1. Introduction

The reconfigurable antennas are considered to be most appropriate candidates for increasing the functionality of wireless communications systems. A system performance can be enhanced by shifting the antenna main beam while maintaining the beam shape, operating frequency, and bandwidth. By utilizing an antenna with reconfigurable radiation patterns, mitigation in the interfering signals and improvement in the desired signal can be attained for enhancement in the wireless communication system. Several designs related to the pattern reconfigurable antenna elements based on metallic patch antenna technology have been proposed and studied [1–7]. These proposed designs exhibit many problems such as large in size, lossy, narrow bandwidth and have low radiation efficiency because of existence of metallic patches. The solution to these problems is to replace the metal patches with a low loss high permittivity dielectric resonator (DR) [8].

The dielectric resonator antenna (DRA) was first introduced by Long et al. in 1983 [9]. The DRA has many advantages such as small size, light weight, design flexibility, low dissipation loss, high radiation efficiency, and ease of excitation as well as wider impedance bandwidth as compared to metallic patch antennas [10, 11]. Different shapes of DRAs such as rectangular, cylindrical, hemispherical, elliptical, pyramidal, and triangular along with various excitation schemes to couple energy into the DR element have been presented in the literature [11–17]. To the authors best knowledge to date, only a few pattern-reconfigurable antennas based on Dielectric Resonator (DR) are reported, where large square ground plane having size of approximately [18] and a circular ground plane having diameter of [19] are presented. Furthermore, in reported articles, multiple coaxial probes are utilized and antenna radiation pattern is reconfigured by exciting one probe at a time while terminating the remaining with matched loads. In our proposed antenna structure, a single coaxial probe is utilized and antenna radiation pattern is reconfigured by electronic switching scheme. Furthermore, antenna ground plane size is significantly reduced having dimensions of .

In this paper, we introduce a novel concept to achieve electronic reconfigurability in radiation pattern of a rectangular dielectric resonator antenna. We will show that with a dual strip excitation scheme along with parasitic strips each having a short/open circuit connection to antenna ground plane can be efficiently utilized to shift antenna radiation pattern at 90° in elevation plane. RF PIN diode switches can be applied on the proposed antenna structure for switching action [20]. Furthermore, the resonance from parasitic strips and the TE₁₁₁ mode of DRA are merged together to achieve wide impedance bandwidth. The organization of the paper is as follows. Antenna design and configuration along with importance of dual strip excitation scheme and parasitic strips are presented in Section 2. In Section 3 simulation results are presented. In Section 4, DRA fabrication along with a comparison between simulation and measured results is given. Finally conclusion and future work to enhance the proposed antenna results are given in Section 5.

2. Proposed Antenna Design and Configuration

2.1. Antenna Design and Configuration

The 3D model of proposed DRA is illustrated in Figure 1. The rectangular DRA has dimensions , , and and dielectric constant . The vertical metallic strip excitation method is chosen to excite the DR due to the ease of implementation, better energy coupling, convenient postmanufacturing trimming, and easy integration with the feed network on the substrate [21]. The two excitation strips are placed in the middle of the DRA side walls in plane having lengths and as shown in Figure 2.

Figure 1 

3D model of proposed DRA structure.

Figure 2 

Proposed DRA structure (𝑋𝑍 plane).

Furthermore, as illustrated in Figure 3 four parasitic strips are placed at each corner of DRA face in plane having length and width . The top view of the antenna structure is illustrated in Figure 4, and the excitation strips are joined together by means of 50 Ω microstrip lines over the substrate. The coaxial probe is located at a distance of from the center of DRA. The four square-shaped grounding pads having dimensions are placed on substrate at a distance of from the each parasitic strip a via hole of radius through the substrate is made in each grounding pad. The four switch locations are also illustrated in Figure 4, where sw₁, sw₂, sw₃ and sw₄ are used to provide a short/open circuit between the corresponding parasitic strip and antenna ground plane. The dimension of substrate backed by a ground plane is 130 × 130 mm² (, where is the free-space wavelength at 1.6 GHz) having thickness of  mm and dielectric constant that is = 4.4.

Figure 3 

Proposed DRA structure (𝑌𝑍 plane).

Figure 4 

Top view of proposed antenna geometry (𝑋𝑌 plane).

Typically, in DRAs impedance bandwidth decreases with the increase in dielectric constant and width/height aspect ratio. Furthermore, the height of DR is used to control the frequency distance between the resonating modes; that is, by increasing the height of the DR the resonating modes can be brought close to each other [22, 23]. In our proposed DRA to achieve wider impedance bandwidth the dielectric constant of DR and aspect ratio is chosen to be 10 and 0.65, respectively. Utilizing the dielectric waveguide model (DWM), the resonance frequency () for rectangular DR can be calculated as follows [24]: where is the free space wavenumber, is the speed of light in vacuum, and , , and are the wave number inside the DR in three directions. The subscripts , , and of denote the number of extremes in the , , and directions, respectively.

For band applications utilizing the above mentioned transcendental equations, the dimensions of dielectric resonator having are found to be  mm,  mm, and  mm, resulting in dominant and higher order mode at 1.85 and 2.05 GHz, respectively.

2.2. Dual Strip for DR Excitation

The DRAs excited by a single point feed usually have a broad side radiation pattern, that is, having a maxima in broadside direction [18, 24, 25]. In our proposed antenna structure, we present a dual strip excitation scheme where excitation strips are placed at two opposite faces of DR and connected together by means of a 50 Ω line. Through this excitation method, we can achieve a radiation pattern having a null in broadside direction with two main lobes having radiation maxima at approximately ±45° in elevation plane. A comparison of antenna radiation pattern with dual and single excitation strip is shown in Figure 5. The initial width and length of excitation strip are kept at  mm and  mm (where and is the free space wavelength at 2.05 GHz), respectively.

Figure 5 

DRA 2D radiation pattern with single and dual strip excitation strips, without parasitic strips at 1.6 GHz, 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=31.2 mm, 𝑊1=5 mm, and 𝑀1=25 mm and 𝑀2=45 mm, 𝑇=3 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm.

2.3. Parasitic Strips for Beam Steering

Next we place four parasitic strips having length and  mm at the corner of DR side wall in plane. Note that initially the length of parasitic strip is kept equal to the height of DR. A connection between the parasitic strip and antenna ground plan is provided by means of a metallic via hole which is placed in grounding pads as shown in Figure 4. It is found that by providing a short circuit between the parasitic strips and antenna ground plane using corresponding switches sw₁ and sw₄, the antenna radiation in regime of positive -axis direction can be suppressed, resulting in antenna main radiation beam to be at in elevation plane. Similarly, using sw₂ and sw₃ to provide a short circuit between the parasitic strip and antenna ground plane antenna radiation in negative -axis direction can be suppressed, resulting in antenna main radiation beam to be at in elevation plane. The switching configuration to achieve a 90° shift in the direction of antenna maximum radiation is shown in Table 1. where, ON state refers to a short circuit between the corresponding parasitic strip and antenna ground plane and OFF state refers to open circuit between the corresponding parasitic strip and antenna ground plane.

Table 1 

Switch configuration for antenna beam steering.

3. Simulation and Optimization

The antenna presented in previous section is simulated and optimized using Ansys high frequency structure simulator (HFSS). The antenna structure is enclosed in a radiation boundary placed at /2 (where is free space wave length at 1.6 GHz) distance away from the antenna structure. The lumped RLC boundary conditions are used to model the switches in the simulation, where the resistance value of 0.001 Kohm and 3 Kohm refers to ON and OFF state, respectively. The simulations are carried out with solution frequency set to 1.6 GHz and fast sweep option is used with 701 points and is kept at 0.02, while number of passes is set to 15 to meet the convergence criteria.

The antenna structure having dual excitation strips and four parasitic strips is optimized to achieve maximum impedance bandwidth and similar radiation patterns for both cases mentioned in Table 1. The initial dimensions are as follows:  mm,  mm,  mm,  mm,  mm,  mm,  mm,  mm,  mm,  mm,  mm,  mm, and  mm.

We studied the antenna radiation pattern in whole frequency band of operation, and we found that the antenna radiation pattern highly deviates in regime of higher order TE₁₁₂ mode. A tedious optimization process is carried out in HFSS to get rid of TE₁₁₂ mode, keeping in mind that this higher order mode that is TE₁₁₂ is highly influenced by the length of excitation strip and can be disposed off by keeping the length of excitation strip below /2 (where is the guided wave length in DR) [25]. It is also observed that by keeping the length of parasitic strip below the half of the height of DR, we can significantly suppress the antenna radiation in unwanted direction. Thus, length of parasitic strip is kept below /2, that is,  mm (where is the height of DR). Furthermore, the feed network consists of 50 Ω microstrip line placed in close proximity of DR over the substrate. We know that these microstrip lines can be used to couple energy to DR directly [26]. This feed network is also optimized to achieve acceptable return loss. Finally, antenna impedance bandwidth of approximately 600 MHz (1.3–1.9 GHz) is achieved with similar radiation pattern characteristics throughout the whole frequency band of operation. The antenna return loss for the both cases mentioned in Table 1 is shown in Figure 6. The final optimized dimensions are as follows:  mm,  mm,  mm,  mm,  mm,  mm,  mm, and  mm and  mm,  mm,  mm,  mm,  mm, and  mm.

Figure 6 

DRA return loss for both cases (Case I and Case II, as mentioned in Table 1), with 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=30 mm, 𝑊1=5.3 mm, 𝐿2=22 mm, 𝑊2=1.5 mm, and 𝑀1=20 mm and 𝑀2=45 mm, 𝑇=2.95 mm, 𝑑sma=44.26 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm.

The parasitic strips height is optimized to achieve maximum impedance bandwidth with similar radiation pattern characteristics. The resonance from parasitic strip and TE₁₁₁ mode of DRA are merged together to achieve wide impedance bandwidth. The optimum height of parasitic strip is found to be  mm with similar radiation pattern characteristics throughout the whole frequency band of operation. The effect of change in parasitic strip height is shown in Figures 7 and 8 for Case I and Case II, respectively.

Figure 7 

Effect of parasitic strip height on DRA impedance bandwidth (Case I, as mentioned in Table 1), with 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=30 mm, 𝑊1=5.3 mm, 𝑊2=1.5 mm, and 𝑀1=20 mm and 𝑀2=45 mm, 𝑇=2.95 mm, 𝑑sma=44.26 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm.

Figure 8 

Effect of parasitic strip height on DRA impedance bandwidth (Case II, as mentioned in Table 1), with 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=30 mm, 𝑊1=5.3 mm, 𝑊2=1.5 mm, and 𝑀1=20 mm and 𝑀2=45 mm, 𝑇=2.95 mm, 𝑑sma=44.26 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm.

The 2D radiation pattern at center frequency of 1.6 GHz for both mentioned cases is shown in Figure 9. The maximum value of antenna radiation beam is found to be at and for Case I and at and for Case II having antenna beam width of for both cases. For a better visualization of beam steering in elevation plane, the antenna 3D radiation pattern is shown in Figure 10. The antenna radiation pattern is tilted towards the negative -axis, when switches are configured as mentioned in Case I (i.e., sw₁ = ON, sw₂ = OFF, sw₃ = OFF, and sw₄ = ON) and tilled towards the positive -axis when switches are configured as mentioned in Case II (i.e., sw₁ = OFF, sw₂ = ON, sw₃ = ON, and sw₄ = OFF). The antenna radiation patterns remain similar with maximum radiation direction changing smoothly from 40° to 45° as the frequency increases. The overall realized gain of DRA is found to be above 4 dB in the whole frequency band of operation (1.3–1.9 GHz) for both cases mentioned in Table 1 and is illustrated in Figure 11.

Figure 9 

2D radiation pattern for both cases mentioned in Table 1, with 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=30 mm, 𝑊1=5.3 mm, 𝐿2=22 mm, 𝑊2=1.5 mm, and 𝑀1=20 mm and 𝑀2=45 mm, 𝑇=2.95 mm, 𝑑sma=44.26 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm at 1.6 GHz. The cross-polar values are too small to be visible in the above scale.

(a)

(b)

Figure 10 

3D radiation pattern for the both cases as mentioned in Table 1, with 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=30 mm, 𝑊1=5.3 mm, 𝐿2=22 mm, 𝑊2=1.5 mm, and 𝑀1=20 mm and 𝑀2=45 mm, 𝑇=2.95 mm, 𝑑sma=44.26 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm. (a) Case I (1.6 GHz) and (b) Case II (1.6 GHz).

Figure 11 

Realized antenna gain for both cases (Case I and Case II, as mentioned in Table 1), with 𝑎=32 mm, 𝑏=29.6 mm, 𝑑=48.85 mm, 𝐿1=30 mm, 𝑊1=5.3 mm, 𝐿2=22 mm, 𝑊2=1.5 mm, and 𝑀1=20 mm and 𝑀2=45 mm, 𝑇=2.95 mm, 𝑑sma=44.26 mm, 𝑠th=1.575 mm, 𝑃𝑑=5 mm, and 𝐿=𝑊=130 mm.

4. Antenna Fabrication and Measurements

The prototype of proposed DRA is shown in Figure 12. The DR having is fixed over the FR4 substrate with ecostock paste having . The excitation and parasitic strips with required dimensions are cut from adhesive tape and placed over the DR faces at appropriate positions. As mentioned earlier, the length of excitation strip is tuned to get rid of mode so that a similar radiation pattern can be attained throughout the whole frequency band of operation. The via holes are drilled in the substrate and filled with silver epoxy to provide a connection between grounding pads and antenna ground plane. The switches are hard-wired for the proof of concept and small pieces of copper wires are used to provide a short circuit between the parasitic strip and corresponding grounding pads, where each grounding pad is connected to antenna ground plane through a metallic via hole.

Figure 12 

Manufactured prototype DRA with 𝑎 = 32 mm, 𝑏=29.5 mm, 𝑑=48.85 mm, 𝜀𝑟=10, 𝐿1=31 mm, 𝑊1=5 mm, 𝐿2=21 mm, 𝑊2=1.5 mm, 𝑀1=22 mm, 𝑀2=33 mm, 𝑠th=1.575 mm, 𝑉rad=0.6 mm, 𝐿=130 mm, 𝑃𝑊=𝑃𝑙=2 mm, and 𝑊=130 mm.

The antenna return loss for the both cases is measured using the network analyzer and is compared with the simulated return loss as shown in Figure 13. The dominant TE₁₁₁ mode is found at approximately 1.79 GHz giving impedance bandwidth of 650 MHz (1.25–1.9 GHz) at center frequency of 1.6 GHz with similar radiation pattern in whole frequency band of operation. A good agreement between the measured, simulated, and predicted (DWM) resonance frequencies is found. The comparison between the theory, simulation, and measured resonance frequencies is shown in Table 2. The difference of impedance bandwidth between measured and simulated results due to fabrication accuracy is found to be approximately 50 MHz. The antenna radiation pattern is measured using the GEOZONDAS time domain measurement setup, and a very reasonable agreement between the simulated and measured results is obtained. The comparison of measured and simulated radiation patterns at 1.3, 1.55, and 1.75 GHz for Case I and Case II mentioned in Table 1 is shown in Figures 14 and 15, respectively.

Table 2 

Comparison of resonant frequencies.

Figure 13 

Comparison between measured and simulated antenna return loss with 𝑎=32 mm, 𝑏=29.5 mm, 𝑑=48.85 mm, 𝜀𝑟=10, 𝐿1=31 mm, 𝑊1=5 mm, 𝐿2=21 mm, 𝑊2=1.5 mm, 𝑀1=22 mm, 𝑀2=33 mm, 𝑠th=1.575 mm, 𝑉rad=0.6 mm, 𝑃𝑊=𝑃𝑙=2 mm, 𝐿 = 130 mm and 𝑊=130 mm.

(a)

(b)

(c)

Figure 14 

Simulated and measured 2D radiation patterns for Case I, with 𝑎=32 mm, 𝑏=29.5 mm, 𝑑=48.85 mm, 𝜀𝑟=10, 𝐿1=31 mm, 𝑊1=5 mm, 𝐿2=21 mm, 𝑊2=1.5 mm, 𝑀1=22 mm, 𝑀2=33 mm, 𝑠th=1.575 mm, 𝑉rad=0.6 mm, 𝑃𝑊=𝑃𝑙=2 mm, 𝐿=130 mm, and 𝑊=130 mm. (a) 1.3 GHz (b) 1.55 GHz (c) 1.75 GHz. The cross-polar values are too small to be visible in the above scale.

(a)

(b)

(c)

Figure 15 

Simulated and measured 2D radiation patterns for Case II, with 𝑎 = 32 mm, 𝑏=29.5 mm, 𝑑=48.85 mm, 𝜀𝑟=10, 𝐿1=31 mm, 𝑊1=5 mm, 𝐿2=21 mm, 𝑊2=1.5 mm, 𝑀1=22 mm, 𝑀2=33 mm, 𝑠th=1.575 mm, 𝑉rad=0.6 mm, 𝑃𝑊=𝑃𝑙=2 mm, 𝐿=130 mm and 𝑊=130 mm. (a) 1.3 GHz (b) 1.55 GHz (c) 1.75 GHz. The cross-polar values are too small to be visible in the above scale.

5. Conclusion

A dielectric resonator antenna with reconfigurable radiation pattern is presented. The DRA consists of rectangular DR with two excitation and four parasitic strips. The DR is mounted over FR4 substrate backed by a ground plane. The DWM theory is utilized to find the appropriate dimensions of DR and dominant TE₁₁₁ mode is generated. It is shown that by utilizing a simple switching scheme for a short/open circuit between the parasitic strips and antenna ground plane the antenna main beam can be reconfigured to an angle of 90° in elevation plane maintaining the similar radiation characteristics as well as impedance bandwidth of 650 MHz (1.25–1.9 GHz).

The proposed antenna is expected to be more efficient than that microstrip patch antenna, especially at millimeter wave frequencies where skin effect is strong. Furthermore, the proposed antenna structure is a potential candidate for the enhancement in array systems to achieve beam steering in azimuth/elevation plane with a wider impedance bandwidth. Effects of physical parameters such as dielectric substrate, excitation strips, parasitic strips, and ground plane will be further studied to improve the radiation pattern reconfigurability. The implementation of RF PIN diodes in proposed antenna structure is underway.

Conflict of Interests

The authors declare that there is no conflict of interests regarding the publication of this paper.

Acknowledgment

This research work is supported by the National Plan for Science and Technology (NPST), Kingdom of Saudi Arabia, under project no. 10-ELE996-02.