Abstract
In this paper, a novel slot antenna array that is based on mirror polarization conversion metasurfaces (MPCM) is proposed. It achieves circular polarization (CP) and effectively reduces the radar cross section (RCS) and increases gain in the entire x-band. This design uses the mirrored composition of the polarization conversion metasurfaces (PCM) on the top surface of the substrate. The MPCM covers a 2 × 2 slot antenna array that is fed with by a sequentially rotating network. The CP radiation is realized by the polarization conversion characteristics of the PCM. At the same time, the reduction of RCS is achieved by 180° (±30°) reflection phase difference between two adjacent PCMs. The improvement in gain is achieved by using a Fabry–Perot cavity, which is constituted by the ground of the antenna and the PCM. Simulated and measured results show that approximately 46.4% of the operating bandwidth is in the range of 7.5–12 GHz (AR < 3 dB) and the gain of the antenna with MPCM is at least 5 dB higher than the reference antenna. Moreover, the monostatic RCS is reduced from 8 to 20 GHz.
1. Introduction
With the rapid development of modern radio technology, great breakthroughs have been made in wireless communication systems, radar navigation, carrier stealth, and other related fields. A multifunction antenna based on electromagnetic metasurfaces can adequately satisfy the requirements of complex systems and concurrent tasks. Therefore, developing a high-gain antenna with a low radar cross section (RCS) that produces circular polarization (CP) is an important subject among experts and scientists [1–8].
A metasurface (MS) is a two-dimensional metamaterial structure with a low profile, low loss, and a simple design. MS has many significant advantages in polarization control, and it has been used in low-RCS antenna as well as RCS reduction design [6–14]. Recently, Akgol et al. proposed a metasurface that uses a polarization converter to convert linear polarization signals to circular polarization signals while having a 3 dB axial ratio bandwidth of 800 MHz [3–5]. In [6–8], the idea of using a partial reflector to achieve a low RCS and high gain was proposed. By adding a resistor to the structure, the energy of the incoming electromagnetic wave can be absorbed to reduce the backward RCS. However, due to the need to weld a large number of resistors, it is difficult to manufacture the antenna in practice. In [10], a new metamaterial surface antenna based on the chessboard polarization conversion metasurface (CPCM) was proposed. Using this approach, a broadband antenna with low RCS, CP, and high gain could be realized by placing a 12 mm MS above the feed antenna. However, the antenna requires a large volume and the axial ratio (AR) bandwidth is narrow. In [11], an MS-based antenna array was proposed, which could realize both circular polarization and RCS reduction, providing a good method for RCS reduction; however, the antenna structure is complex and the impedance bandwidth is narrow. Although the method proposed above has made significant progress, the application of MS experiences considerable shortcomings. In the state-of-the-art designs, low-profile antennas that simultaneously achieve wideband RCS reduction and wideband CP remain a challenge.
To further realize the integration of multiple performances on one antenna, this paper proposes a broadband low-RCS, low-profile CP, high-gain antenna based on mirror supersurface. The AR bandwidth is expanded by means of sequential rotation feeding [15, 16]. Circular polarization, RCS reduction, and gain enhancement can be realized in the entire x-band. The antenna array achieves a larger RCS reduction and AR bandwidth than the CP slot antenna that is described in published literature. Compared with the low-RCS Fabry–Perot resonant cavity that is based on MS, the proposed antenna has an important advantage. It is low profile and without air gap, which means that it can be applied in many fields.
2. Design of the Mirror Polarization Conversion Metamaterials
2.1. Structure of the Antenna
Figures 1(a)–1(c) illustrate the structure of the antenna, which is composed of an MPCM and a slot antenna array (as the source antenna). The MPMS consists of four parts: Part I is an MPCM, parts II and IV are its mirror images, and part III is the mirror image of part II. Figure 2(a) shows an enlarged view of the PCM with a linear polarization (LP) slot antenna source. The PCM is composed of 4 × 4 square cells with different corners truncated on both sides. In order to expand the AR bandwidth, the feed antenna is supplied data from the network as it rotates in sequence. The phase distribution is 0°, 90°, 180°, and 270°, as shown in Figure 1(c). The substrate utilized is Rogers RO4003C with = 3.55 and a loss tangent of δ = 0.0027. The optimized widths are as follows: = 0.18 mm, , , , , , , , and and r = 3.5 mm.
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Each PCM performs three functions. First, when an LP wave passes through the PCM, it is converted into CP wave. Second, the PCM can achieve reflective wave wideband polarization conversion. Third, an increase in the gain is realized by the Fabry–Perot cavity that is formed by the slot array antenna and the PCM. Obviously, it is very difficult to adjust these three functions at the same time. Therefore, the objective of this research is to balance these features. The general procedure for the design of the PCM can be summarized as follows:(1)The unit cell PCM, covering the operation band (8–12 GHz) of the antenna, is designed to obtain the initial parameters(2)The copolarization and cross-polarization transmission coefficients are modified by optimizing the diagonal corner truncated on both sides of the unit cell. This will convert the LP wave into the CP wave(3)Optimize the x-component and y-component of the reflection coefficient to meet the Fabry–Perot resonance condition of CP(4)Optimize the polarization conversion unit cell with flooring (Figure 3) to realize broadband polarization conversion(5)Repeat steps 2–4 until the process is complete
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Following the design steps above, the final parameters of the unit cell are as follows: a = 4.6 mm, d1 = 1.2 mm, d2 = 2.8 mm, c = 0.4 mm, b = 25 mm, Ws = 2.2, Ls = 14.4 mm, L1 = 3.9 mm, h0 = 0.508 mm, and h1 = 2 mm.
2.2. Analysis of the Circular Polarization
One way to describe an array antenna is that it consists of four PCM-based CP antennas, which are arranged in 90° rotations. An equivalent circuit can be used to explain the formation of CP. In Figure 2(a), the structure within the area enclosed by the red square can be considered a new unit. When the PCM is placed on the slot antenna, which is LP along the y-axis, an E-field is generated along the y-axis. The E-field is broken down into orthogonal components and , as shown in Figure 2(c). If the diagonal of the unit is not truncated, due to the symmetry of the structure, as shown in Figure 2(b), the orthogonal components and constitute an RLC circuit with an impedance given by [17]where L and R are the inductance and resistance of the patch and C is the capacitance generated by the gap between nearby and related units. When the diagonal corners are truncated as shown in Figure 2(c), the perpendicular E-field components that are broken down by the PCM will cause two different impedances, and . The expression for the two impedances is shown in the following equations:
The truncation corner increases the gaps between nearby and related patches, thereby increasing the value of and decreasing the value of . Thus, the phase difference between and can be changed by modifying the value of the truncated corners on both sides. If the PCM is designed such that = and = 90°, then and = 90°.
Figure 4(a) presents a comparison of the reflection coefficients for slot antennas both with and without PCM. As shown in Figure 4(b), the axial ratio (AR < 3 dB) bandwidth of the slot antenna with PCM ranges from 9.2 to 10 GHz. The results show that PCM can have the ability to transform the LP wave into a CP wave. In addition, the gain has been improved. The reasons for this improvement are explained in the following section.
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2.3. Mechanism of High-Gain Performance
A metamaterial surface (MS) is essentially the surface distribution of a small electrical scatterer [18]. Due to its planar structure, it can be easily combined with a planar antenna to form a resonant cavity. As shown in Figure 2(a), when the spacing between the reflective floor and the MS meets the resonance condition, the electromagnetic wave is constantly reflected back and forth in the resonant cavity. Also, the electromagnetic wave transmitted through the MS can be superimposed with the phase each time. Thus, the gain and beam width of the antenna are improved, and the radiation energy can be directed by adjusting the height between the two plates. According to the analysis in the literature [19], two resonant modes (TM10 mode and antiphase TM20 mode) are simultaneously excited to load the metamaterial element of the slot antenna. This in turn leads to the low-profile wideband LP antenna. Due to the distribution of the truncated diagonal corner, the impedance matching bandwidth is improved. However, the gain is less affected. Figures 5(a) and 5(b) show the electric field distribution at two distinct frequencies (8.6 and 10.5 GHz), verifying its two resonant modes: TM10 mode and TM20 inversion mode.
2.4. Analysis of RCS Reduction
As shown in Figure 3, the MPCM patch can be regarded as a checkerboard surface that consists of a PCM array and mirror PCM array. A phase difference of 180° is created between each part, so that the reflected waves for each will cancel each other out. Consequently, the total RCS is significantly reduced. In order to study the RCS reduction characteristics of the array antenna, the polarization conversion characteristics of the PCM element need to be studied. When the incident wave impinges along the z-direction, the unit cell can be considered a reflective polarization converter. As shown in Figure 3, using the linear x-polarized normal-impinging wave as an example, the reflected electric fields and along the x-axis and the y-axis are generated. Definitions (copolarization) and (cross-polarization) represent the reflection ratio in the x-x and x-y directions, respectively. The polarization conversion ratio (PCR) is defined as , representing the polarization conversion characteristics for the PCM cell. The PCM unit cell has been simulated in the software Ansys HFSS v13.0, using the Floquet port and master/slave boundary conditions. The results presented in Figure 6(a) show that the copolarization of the reflection coefficient loses its dominant position within 8–20 GHz, whereas the cross-polarization becomes the dominant component. The PCR value is greater than 0.7 between 8 and 20 GHz, which means that most of the energy is redirected in the orthogonal direction. Figure 6(b) presents the cross-polarization reflection phase difference between the PCM infinite period unit and its mirror. In the working frequency band, the reflected wave of PCM has the same amplitude as that of the corresponding mirror image, with a phase difference of 180°. The reflected waves from each part cancel each other out, decreasing the RCS. Due to the symmetrical structure of the MPCM, the reduction in RCS of the y-polarized incident wave is similar. As a result, the mirror chessboard PCM structure is chosen due to its broadband, low-RCS features.
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Considering the arrangement of MPCM, the corresponding slot antennas under each PCM should not be configured such that they are in the same direction, and they should not be excited by the same phase. Therefore, a CP slot array antenna based on a sequential rotational feed network is selected as the source antenna. Because the reference antenna also utilizes circular polarization radiation, it may seem unnecessary to use MPCM for the same purpose. However, antennas using MPCM have a wider AR bandwidth because a superior polarization purity can be obtained by replacing four LP components with four CP components. In addition, gain can be increased at operating frequencies, so the radiation performance and scattering characteristics are balanced by the combination of the arrangement of the MPCM and the radiation source.
3. Simulation and Measurement Comparison of Both Antennas
In order to verify the correctness of simulation and analysis, an array sample with a size of 50 × 50 × 2.5 mm2 was created and measured. The spacing of each slot is 20.7 mm, approximately 0.6 (relative to the operating frequency of 10 GHz). The radiation patterns of the antenna are measured in a microwave chamber without reflection, as shown in Figure 7(a). The reflection coefficient of the antenna was measured using an Agilent N5247A vector network analyzer.
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Figure 7(b) shows the reflection coefficient results, both with and without the MCPM antenna simulation and measurement. Although there exists some deviation, the overall results are consistent. The measurement and simulation AR bandwidth for the two antennas are shown in Figure 7(c). The measured CP operating bandwidth is 8−12 GHz (40%), whereas the simulated CP operating bandwidth is 7.5–12 GHz (46.1%). The results indicate that the AR bandwidth can be extended by using the MPCM. Figure 7(d) shows the gain versus frequency plot for the simulation and measurement of the antenna. The gain of an antenna with MPCM is higher from 7–11.2 GHz, compared to that of an antenna without MPCM. The measured 3 db gain is in the 7.7–10.5 GHz band (30.7%), whereas the simulated one is in the 7.9–10.8 GHz band (31%). The main cause for the offset in frequency is attributed to both manufacturing error and error in measurement. Figures 8(a) and 8(b) depict the measurement and simulation patterns of the two antennas at 9.3 GHz. The difference between simulation and measurement is due to the measurement environment, manufacturing tolerances, and assembly errors. The test and simulation results in Figures 7 and 8 show good consistency overall, verifying that the use of MPCM improves the radiation performance.
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The 3D bistatic scattering field at 10 GHz under normal incidence, both with and without the MPCM antenna, is shown in Figures 9(b) and 9(c). The normal directional energy of the scattered wave is redistributed in four directions (, ) = (45°, 46°), (135°, 46°), (225°, 46°), and (315°, 46°). The frequency response of the measured and simulated monostatic RCS in the x-polarized normal-impinging plane wave can be seen in Figure 9(a). The implemented RCS reduced frequency bandwidth is from 8 to 20 GHz, and the RCS reduced band is consistent with the frequency band for PCR values greater than 0.7. In addition, the maximum RCS reduction value is at least 14 dB. Since the MPCM structure is symmetrical, the same effect is produced by a normal-impinging plane wave with y-polarization. Some differences can be observed between the simulation and the measurement; this occurs primarily because of tolerance during fabrication, measurement deviations, and environmental factors, all affecting the measured RCS results.
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4. Conclusions
In this paper, a novel antenna is proposed. It features a broadband CP, high gain, and low RCS using MPCM. The RCS range is between 8 and 20 GHz and is lower than the reference antenna. Furthermore, the gain is increased in the operating bandwidth and the AR bandwidth is also broadened. The simulated and measured results show that there is an improvement in the scattering and radiation characteristics of the MPCM antenna. Therefore, the design presented in this research provides a new approach for implementing broadband CP, high-gain, and low-RCS antennas. This has potential for use in several applications, most notably in stealth platforms.
Data Availability
The data used to support the findings of this study are included within the article.
Conflicts of Interest
The authors declare that there are no conflicts of interest regarding the publication of this paper.
Acknowledgments
This research was partially supported by the Major Project of Provincial Natural Science Research of the University of Anhui Province of China (Grant nos. KJ2018ZD046 and 2017sxzx40), the Science and Technology Project of Anhui Province (Grant no. 1708085QF150), and the National Natural Science Foundation of China (Grant nos. 61801163 and 61871001).