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Journal of Electrical and Computer Engineering
Volume 2012, Article ID 218930, 9 pages
Research Article

Ranging Performance of the IEEE 802.15.4a UWB Standard under FCC/CEPT Regulations

1Signal Processing and Speech Communication Laboratory, Graz University of Technology 8010 Graz, Austria
2Communication Technology Laboratory, Swiss Federal Institute of Technology (ETH), 8092 Zurich, Switzerland
3CISC Semiconductor, Design and Consulting GmbH, 9020 Klagenfurt, Austria

Received 25 October 2011; Accepted 29 January 2012

Academic Editor: Hsien-Chin Chiu

Copyright © 2012 Thomas Gigl et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.


The IEEE 802.15.4a standard for wireless sensor networks is designed for high-accuracy ranging using ultra-wideband (UWB) signals. It supports coherent and noncoherent (energy detector) receivers, thus the performance-complexity-tradeoff can be decided by the implementer. In this paper, the maximum operating range and the maximum allowed pathloss are analyzed for ranging and both receiver types, under FCC/CEPT regulations. The analysis is based on the receiver working points and a link budget calculation assuming a frees-pace pathloss model. It takes into consideration the parameters of the preamble, which influence the transmit power allowed by the regulators. The best performance is achieved with the code sequences having the longest pulse spacing. Coherent receivers can achieve a maximum operating range up to several thousand meters and energy detectors up to several hundred meters.

1. Introduction

Real-time locating systems (RTLSs) and sensor networks are challenging topics for research and development. Novel applications, such as the tracking of fire fighters in emergencies [1, 2] and the tagging of cars in the manufacturing process, need very precise and reliable localization in multipath intensive environments. Common approaches as, for example, the global positioning system (GPS) or WLAN fail in such situations, because the signals are unable to penetrate the roof and the walls and/or they get disturbed by multipath propagation. Thus, researchers have focused on new radio frequency (RF) technologies in recent years, in particular ultra-wideband (UWB). UWB shows robustness against multipath interference and allows for highly accurate positioning [39].

IEEE 802.15.4a [10] is a standard for wireless sensor networks with submeter ranging accuracy in indoor environments. The physical layer is designed for bidirectional communications using amplitude and position modulated, bandpass-type UWB signals. It thus supports high-complexity coherent receivers and low-complexity energy detectors. A two-way time of arrival ranging scheme is proposed. The first part of each transmission is a pulse sequence with known codes, called the preamble, which is used for signal detection, synchronization, and the estimation of the channel impulse response (CIR) that is needed to obtain the time-of-arrival. The preamble sequences show perfect autocorrelation properties for both receiver types [11]. Energy detectors promise low cost and low power consumption, but a performance loss occurs [12] and more vulnerability is unavoidable with respect to interfering signals [13]. Sub-Nyquist-rate sampling can further reduce the complexity, but also the ranging performance [7, 14].

The IEEE 802.15.4a standard has a large number of system parameters that influence the achievable ranging performance [1517]. Based on the results in [18], the maximal allowed transmit power is analyzed in this paper taking into account the regulations of the Federal Communication Commission (FCC) [19] and the Conférence Européenne des Postes et Télécommunications (CEPT) [20]. The system performance is evaluated with respect to the achievable operating range and the maximum allowed pathloss for a coherent receiver and an energy detector. Our analysis shows the parameter settings and frequency channel selection for maximizing the performance.

The paper is organized as follows. Section 2 presents the signal models according to the 802.15.4a standard, a signal-to-noise ratio (SNR) analysis, and the performance metric definition. The FCC and CEPT regulations are discussed in Section 3, and the link budget is introduced in Section 4. This is followed by performance results and conclusions in Sections 5 and 6.

2. Problem Statement

An (indoor) ranging system needs to find the line-of-sight (LOS) component in the channel response, because the detection of a reflection or a noise component may lead to very large errors. Thus, the ranging performance can be characterized by the quality of the channel estimation at the receiver output. An appropriate performance metric is the receiver output SNR of the LOS component (LSNR), as it correlates strongly with the ranging performance [21]. It is defined as ||𝑦LSNR=𝑠𝑛LOS||2[𝑛]var,(1) where 𝑦𝑠[𝑛LOS] is the LOS sample 𝑛LOS of the receiver output and var{[𝑛]} is the noise variance of the estimated channel response [𝑛].

To study the operating range, it is necessary to relate the output SNR to the input SNR of the receiver. The input SNR is given by the transmit power regulations, the link budget, and the channel. The output SNR furthermore depends on the receiver structure, the hardware components, and—in case of a noncoherent receiver—also on the preamble parameters, as analyzed below. The input SNR is defined by the energy of the despread LOS component over the noise spectral density 𝐸LOS/𝑁0 with 𝐸LOS=𝑀1𝐸(1)LOS, where 𝐸(1)LOS is the received energy for the LOS component of a single pulse and 𝑀1 is the number of transmitted pulses.

2.1. Signal Models

This section introduces the signal models for the IEEE 802.15.4a physical layer and the receivers.

2.1.1. IEEE 802.15.4a

The most important signal part for ranging is the preamble. It employs a code sequence 𝐜𝑠 of length 𝑁𝑠=31 or 127 [10] that consists of ternary elements {1,0,1}. The preamble code vector 𝐜sp is created as 𝐜sp=𝟏𝑁pr𝐜𝑠𝜹𝐿=𝐜𝜹𝐿,(2) where denotes the Kronecker product, 𝜹𝐿 is a unit vector with a one at the first position and length 𝐿 to extend the spacing between the preamble chips, and 𝟏𝑁pr denotes a vector of ones to repeat the preamble sequence 𝑁pr times. The vector 𝐜 is the periodically repeated preamble code. The transmitted signal 𝑠(𝑡) is defined as 𝑠(𝑡)=𝐸𝑝𝑀1𝑚=0𝑐𝑚𝑤𝑡𝑚𝐿𝑇chip𝑒𝑗𝜔𝑐𝑡=𝐸𝑝𝑀1𝑚=0𝑐𝑚𝑤𝑡𝑚𝐿𝑇chip,(3) where 𝐸𝑝 is the energy per pulse, 𝑐𝑚 is the 𝑚-th element of 𝐜, 𝑤(𝑡) is the energy-normalized pulse shape, 𝑀 is the number of code elements in the preamble, 𝜔𝑐 is the carrier frequency, 𝑇chip is the chip duration, and 𝑤(𝑡) is the upconverted pulse assuming the carrier and the pulse are phase synchronous.

Table 1 shows the timing characteristics of the preamble, where 𝑇pr is the total duration of the preamble, PRF is the peak pulse repetition frequency, MRF is the mean pulse repetition frequency, and 𝑁1ms is the number of preamble sequences within 1 ms. ERF is the effective pulse repetition frequency according to the regulations (see Section 3).

Table 1: Preamble characteristic.

The transmitted signal (3) is sent over a multipath channel with channel impulse response 𝑐(𝑡), where also the effects of the antenna are contained for simplicity. Furthermore, 𝑐(𝑡) is assumed to be constant during 𝑇pr. Thus, the analog received signal is obtained from 𝑟𝑎(𝑡)=𝑠(𝑡)𝑐(𝑡)+𝜈(𝑡),(4) where 𝜈(𝑡) is modeled as additive white Gaussian noise and is the convolution. Next, the receiver architectures are described.

2.1.2. Coherent Receiver

Figure 1 shows the system model of the coherent receiver. The signal is received by a UWB antenna and filtered by the transmit pulse shape 𝑤(𝑡). Thus, a matched filtering to the pulse shape is applied. The signal is converted to complex baseband using the Hilbert transform hilb(𝑡) and carrier demodulation by the estimated frequency 𝜔𝑐. The complex baseband signal is given by 𝑟𝑏𝑟(𝑡)=𝑎𝑤(𝑡)(𝑡)hilb𝑒(𝑡)𝑗𝑤𝑐𝑡+𝜑,(5) where 𝜑 is the unknown carrier phase. Assuming synchronization and known carrier frequency, an estimated sampled channel response is obtained after despreading, [𝑛]=𝑀1[𝑛]+𝑁pr1𝑁𝑞=0𝑠1𝑚=0𝑐𝑚𝜈𝑏𝑛+𝑚+𝑞𝑁𝑠𝐿𝑁chip,(6) because the preamble codes have perfect circular autocorrelation properties, thus interpulse interference (IPI) is canceled. (For this assumption it is necessary that the maximal excess delay 𝜏max𝐿𝑁𝑠𝑇chip=𝑇𝑠, where 𝑇𝑠 is the period of the spread preamble sequence. The IEEE 802.15.4a standard has a 𝑇𝑠1μs, which is usually sufficient for IPI free processing in indoor environments.) The despreading is first performed sequencewise (𝑚) and then over the sequence repetitions (𝑞). Since 𝑐2𝑚=1 for the nonzero code elements, it follows that 𝑞 and 𝑚 simply the number of nonzero code elements in the preamble 𝑀1=((𝑁𝑠+1)/2)𝑁pr, that is, the number of transmitted pulses. The number of samples within a chip is defined by 𝑁chip=𝑇chip/𝑇. The noise 𝜈𝑏[𝑛] is the band-limited input noise 𝜈𝑏[𝑛]=𝜈[𝑛]𝑤[𝑛] in complex baseband. A detailed derivation of the equations can be found in [15].

Figure 1: Coherent receiver.

This receiver architecture needs high sampling rates according to the Nyquist theorem. Another disadvantage of this concept is the required synchronization of the carrier frequency and phase, which is critical for its performance. The energy detector is based on a different method for the downconversion that prevents these two problems. Thus, a low-complexity solution is obtained.

2.1.3. Energy Detector

The energy detector works as shown in Figure 2. The signal is again received by a UWB antenna and filtered by a bandpass filter, which ideally is matched to the pulse shape. Next, the signal is squared and integrated for short-time windows 𝑇𝐼. The length of 𝑇𝐼 also defines the sampling period. It causes a mean absolute error (MAE) of ranging greater or equal 𝑇𝐼/4 [7], which limits 𝑇𝐼 to a few ns for highly accurate ranging. The signal model after sampling is given by 𝑥[𝑛]=(𝑛+1)𝑇𝐼𝑛𝑇𝐼𝑟𝑎(𝑡)𝑤(𝑡)2=𝑑𝑡(𝑛+1)𝑇𝐼𝑛𝑇𝐼𝑀1𝑚=0𝑐𝑚𝑔𝑡𝑚𝐿𝑇chip+𝜈𝑓(𝑡)2𝑑𝑡,(7) where 𝜈𝑓(𝑡) is the passband filtered noise and the channel response 𝑔(𝑡)=𝐸𝑝𝜙𝑤(𝑡)𝑐(𝑡+𝑚𝐿𝑇chip) and 𝜙𝑤(𝑡) is the autocorrelation function of 𝑤(𝑡). The estimated channel response 𝑦[𝑛] is obtained by despreading 𝑥[𝑛], 𝑦[𝑛]=𝑁pr1𝑁𝑞=0𝑠1𝑖=0̃𝑐𝑖𝑥𝑁𝑛+𝑖𝐿chip+𝑞𝑁𝑠𝐿𝑁chip=𝑦𝑠𝑠[𝑛]+𝑦𝑠𝜈[𝑛]+𝑦𝜈𝜈[𝑛],(8) where 𝑁chip=𝑇chip/𝑇𝐼. The code despreading is performed sequencewise with 𝑞 and 𝑖 with the despreading code ̃𝑐𝑖. In contrast to the coherent receiver, the noncoherent receiver uses a different despreading code ̃𝐜 than the spreading code to obtain perfect circular correlation properties for the squared sequences. This code is created by squaring 𝐜 and setting all zeros to −1 [11]. The output of the energy detector comprises a signal-by-signal term 𝑦𝑠𝑠[𝑛], a linear signal-by-noise term 𝑦𝑠𝜈[𝑛], and a quadratic noise-by-noise term 𝑦𝜈𝜈[𝑛]. The code correlation can completely cancel the IPI in the signal term 𝑦𝑠𝑠[𝑛] but not for the cross-term 𝑦𝑠𝜈[𝑛]. A longer pulse spacing leads to less IPI such that it becomes negligible in indoor environments with a spacing of 𝐿16 [15, 16]. The full derivation of the equations can also be found in these references.

Figure 2: Energy detector.
2.2. Input-to-Output SNR Relation

The input-to-output SNR relation for the coherent receiver is given by [15] LSNRCR=𝐸LOS𝑁0.(9) For the energy detector, the relation is given by [15, 16] LSNRED=2𝐸LOS/𝑁024𝐸LOS/𝑁0+𝑁𝑠𝑁pr𝑇𝐼𝑊RRC,(10) where 𝑊RRC is an equivalent bandwidth defined as 𝑊RRC=𝜙2𝑤(𝜇)𝑑𝜇. The first and second terms of the denominator correspond to the variance of the linear and the quadratic noise terms, respectively. The quadratic noise term depends on the receiver parameters that can be combined to the noise dimensionality ND=𝑁𝑠𝑁pr𝑇𝐼𝑊RRC [13]. For practical values of ND, the output SNR is proportional to (𝐸LOS/𝑁0)2, while it shows a linear relation to 𝐸LOS/𝑁0 for the coherent receiver.

Figure 3 shows the relation of the detector input SNR 𝐸LOS/𝑁0 and the output LSNR based on (9) and (10). The specific curves for the energy detector (ED) are obtained by increasing ND by factors of four. The depicted curves correspond to 𝑁pr=[16,64,256,1024,4096], 𝑁𝑠=31, 𝑇𝐼=2ns, and 𝑊RRC=1GHz. Note that 𝑁pr=256 is not included in the standard. The curves are separated if the quadratic noise term dominates and they merge if the linear noise term is dominant. Increasing 𝐸LOS/𝑁0 by 6dB leads to LSNR +12dB in the quadratic part and to +6dB in the linear one. The horizontal line illustrates the LSNR at the working point LSNRWP=12dB for the energy detector (cf. [16]). At this working point, 80% of the range estimates are within 1m. The coherent receiver shows a working point LSNRWP=9dB (cf. [21]). Both working points have been determined by extensive simulations (see [15]). It can be seen for the energy detector that 3dB more 𝐸LOS/𝑁0 are required when 𝑁pr, the number of sequence repetitions, is increased by a factor of four. Note that the quadratic noise term dominates at this working point. As observable from (9), the LSNR for the coherent receiver is independent of the number of the pulses. It depends only on the transmitted energy. In other words, it does not matter if this energy is transmitted in one pulse or in a sequence of pulses. The coherent receiver shows an advantage ≥11 dB in the working point in comparison to the noncoherent receiver.

Figure 3: Relation between input SNR (𝐸LOS/𝑁0) and output SNR (LSNR) with respect to the noise dimensionality ND=𝑁𝑠𝑁pr𝑇𝐼𝑊RRC for the energy detector (ED) and the coherent receiver (CR). The fixed parameters are 𝑁𝑠=31, 𝑇𝐼=2.0032ns, and 𝑊RRC=1GHz.
2.3. Maximal Operating Distance

As 𝑁0 is constant in the scenario, 𝐸LOS/𝑁0 for the maximal operating distance 𝑑max is obtained from the well-known pathloss model 𝐸LOS𝑁0𝑑maxdB=𝐸LOS𝑁0𝑑0dB𝑑10𝜂logmax𝑑0,(11) where 𝜂 is the pathloss exponent and 𝑑0 is a reference distance. The maximal operating distance for the coherent receiver is obtained from (9) and (11): 𝑑max=𝐸LOS/𝑁0(1m)𝐸LOS/𝑁0𝑑max1/𝜂=𝐸LOS/𝑁0(1m)LSNRWP1/𝜂,(12) where the reference distance 𝑑0 is assumed to be 1m. It follows for the energy detector 𝑑max=𝐸LOS/𝑁0(1m)LSNRWP+LSNRWPLSNRWP+ND/21/𝜂(13) using (10) in (11).

2.4. Maximal Allowed Pathloss PLmax

A more general look at the achievable range is given by the maximal allowed pathloss, which is independent of the channel model, fading margins, or implementation losses. The pathloss model can be rewritten with (9) to obtain PLmax for the coherent receiver PLmax,dB=𝐸LOS𝑁0(1m)dBLSNRWP,dB,(14) where 𝐸LOS is the energy of the received LOS component at 1 m, which does not take fading margins or implementation losses into account (see Table 2).

Table 2: Link budget.

For the energy detector follows, using (10) and (11), PLmax,dB=𝐸LOS𝑁0(1m)dB10logLSNRWP+LSNRWPLSNRWP+ND2.(15)𝐸LOS/𝑁0(1m) and 𝐸LOS/𝑁0(1m) are defined by the transmitted preamble energy 𝐸pr (see Section 3) and the link budget (see Section 4).

3. FCC Regulations

In this section, the maximal allowed transmit power is calculated with respect to the FCC regulations [19]. In principle, the same regulations have been adopted by the CEPT in Europe for the band between 6and8.5GHz [20]. In the band between 3.1 and 4.8GHz, the CEPT requires detect and avoid (DAA) or low duty cycle (LDC) mitigation additionally, which does not influence this analysis. This analysis is done in accordance to [18].

The FCC constraints essentially consist of an average and a peak power limit. In any band of bandwidth 𝐵av=1MHz, the average transmit power is limited to 𝑃FCCav=41.3 dBm for an averaging window of 𝑇av=1ms. The peak power within the bandwidth 𝐵pk=50MHz is restricted to 𝑃FCCpk=0 dBm. Both peak and average transmit power are defined by the equivalent isotropically radiated power (EIRP).

The 802.15.4a preamble is a sequence of nonuniformly spaced pulses whose polarities are chosen pseudorandomly by the codes. According to [18], its average and peak power are determined by ERF and PRF, respectively. The pulse energy spectral density (ESD) 𝐸𝑝,av|𝑊(𝑓𝑐)|2 for the average power limit is given by 𝐸𝑝,av||𝑊𝑓𝑐||2=𝑇av𝑃FCCav2𝐵av1,ERF𝑇av,𝑃FCCav2𝐵av1ERF,ERF𝑇av,(16) where 𝐸𝑝,av is the pulse energy limited by the average power limit and 𝑊(𝑓𝑐) is the spectrum of the normalized pulse 𝑤(𝑡) at the center frequency 𝑓𝑐. ERF is defined as 𝑇ERF=MRFpr𝑇av=𝑀1𝑇av𝑇pr<𝑇avMRF𝑇pr𝑇av(17) where 𝑀1=(𝑁𝑠+1)/2 is the number of code elements not equal to zero. ERF is the compressed MRF due to stretching the preamble over the averaging time 𝑇av=1 ms (FCC). If 𝑇pr is greater than 𝑇av, then it is MRF. The mean power is limited by the number of pulses within 1 ms (𝑁1ms) (see Table 1).

The peak power limit is defined by the PRF, where the sequenced pulses within an observation window 1/𝐵pk=20ns are added. The ESD 𝐸𝑝,𝑝|𝑊(𝑓𝑐)|2 for the peak power limit is obtained by 𝐸𝑝,𝑝||𝑊𝑓𝑐||2=𝑃FCCpk9𝐵2pk3,PRF2𝐵pk,𝑃FCCpk4PRF23,PRF2𝐵pk,(18) where 𝐸p,p is the pulse energy limited by the peak power limit.

The maximal FCC compliant pulse ESD with respect to peak and average power is shown in Figure 4. To find the active ESD for a specific preamble, the smaller value between ESDpk at PRF and ESDav at ERF is considered. The peak power limit for the short preamble symbols with 𝐿=16 and 𝐿=64 is the same, while, for the long preamble sequences it is lower due to higher PRF. It can be observed that only the preamble sequences with 𝑁pr=16 are peak power limited. However, it is reported in [22] that the supply voltage limits the transmit power in low-data-rate systems and the peak power limit cannot be exploited for low supply voltages.

Figure 4: Pulse energy spectral density.

Assuming a pulse with rectangular spectrum, the energy per pulse 𝐸𝑝=2𝐵𝐸𝑝|𝑊(𝑓𝑐)|2, where 𝐵 is the pulse bandwidth. Thus, the achievable preamble SNR can be calculated as shown in Figure 5. At 𝑁pr=16, all preamble codes are limited by the peak power limit. The long preamble symbols contain approximately. 2 dB more energy in four times more pulses. Increasing the number of pulses does not necessarily lead to a preamble energy improvement, if 𝑇pr1 ms, because the long preamble symbols and the short codes with spreading 𝐿=16 are mean power limited between 𝑁pr=64 and 1024. 𝑁pr=4096 leads to an improvement, because 𝑇pr>4 ms, which means the preamble is more than four times longer than 𝑇av. The short preamble codes with spreading 64 imply a four times longer preamble in contrast to the others, thus a gain of up to 6 dB can be achieved.

Figure 5: Achievable preamble energies.

4. Link Budget

As mentioned before, 𝐸LOS/𝑁0(1m) is the input SNR of the receiver at 1m which depends on the link budget. Table 2 shows an example link budget calculation for 802.15.4a channel (ch) 3, using 𝑁pr=1024, 𝑁𝑠=31, 𝐿=16, 𝑓𝑐=4.4928GHz, and a bandwidth of 499.2MHz. In that case, the average power limit of the FCC regulations applies and 𝐸𝑝 is calculated from (16), where also the antenna gain is included. 𝐸𝑝 is limited for 1006 sequences due to averaging over 1ms (see Table 1). The preamble energy 𝐸pr=𝐸𝑝𝑀1, the free space loss 𝐿fs at 1m is given by 45.5dB using Friis’ equation [23], and the receiver antenna gain 𝐺RX is defined by 0dBi. These values yield the received preamble energy without multipath components, meaning the energy of the line of sight component 𝐸LOS at 1 m. Assuming the input structure of the receiver is linear, the noise spectral density is given by 𝑁0=𝑘𝑇0𝐹 [23], where the Boltzmann constant 𝑘=1.38×1023 Joule/Kelvin [J/K], the temperature of the environment 𝑇0=293K, and the noise figure of the receiver input structure 𝐹=5dB (cf. [24]). Implementation losses of 4dB and a LOS fading margin of 3dB are assumed. Thus, 𝐸LOS/𝑁0(1m) is obtained and can be used to calculate the maximal operating range according to (11) and (13).

5. Results

The maximal operating distance and the maximal acceptable pathloss are analyzed in this section. The maximal operating distance is based on the free-space link budget, because the LOS component is needed for accurate ranging. The maximal acceptable pathloss is shown as a more general value, which allows the implementer to analyze the effect of specific channel models, for example, NLOS scenarios, or specific system parameters, for example, lower noise figures.

5.1. Effect of Codes

Figure 6 shows the maximal operating distance 𝑑max and the maximal allowed pathloss PLmax with respect to the length 𝑁pr of the preamble sequences. The coherent receiver is directly proportional to 𝐸𝑝/𝑁0, which means that the operating distance is related to the preamble energy discussed in Figure 5. By contrast, the ED suffers from noncoherent combining losses, thus the noise dimensionality including the number of transmitted pulses is important for the final performance (see (13)). A change of 𝐸𝑝/𝑁0 without changing the noise dimensionality, for example, using a different noise figure or pulse energy, leads to a shift of the curves along the dashed lines, but the shape of the curves does not change. Thus, the general conclusions are still valid, while 𝑑max and PLmax need to be recalculated according to the new link budget.

Figure 6: Code sequence analysis on the maximal operating distance and the maximal allowed pathloss, where the solid lines correspond to the energy detector, the dashed-dotted lines belong to the coherent receiver, and the dashed lines indicate constant noise dimensionality for the energy detector. The values are given for 𝑁pr=[16,64,256,1024,4096] (from left to right).

In the overall performance there is a big gap between the CR and the ED. The CR achieves a maximal operating distance up to several thousand meters and the ED achieves only several hundred meters. However, an operating distance of several hundred meters is usually sufficient for (low-complexity) indoor localization systems and sensor networks.

As expected from Section 3, the best performance is achieved by the short preamble with long spreading (𝐿=64) which has the highest transmitted energy. A maximal operating distance of approximately 6000m (PLmax82dB) is achieved by the CR, and 430 m (PLmax60dB) is achieved by the ED. The CR reaches approximately half of that distance (3000 m; PLmax76dB) for the other two codes. As mentioned before, the energy detector shows a more specific behavior, which is discussed in detail in the rest of this paragraph. As observable, the increasing of 𝑁pr does not necessarily lead to a better performance. A performance degradation is seen at around 𝐸LOS/𝑁0(𝑑0)79dB for increasing numbers of transmitted pulses due to constant transmitted energy (cf. Figure 5). This effect also harms the performance of the long preamble codes (𝑁𝑠=127) significantly and leads to the lowest performance achieved. The performance of the short preamble sequences (𝑁𝑠=31) with spreading 𝐿=16 and the long preamble sequences (𝑁𝑠=127) is best at 𝑁pr=64, where a distance of 300m (PLmax56dB) and 200 m (PLmax53dB) is reached, respectively. The preamble sequences with 𝐿=64 show a local optimum for 𝑁pr=64 with the same performance as the 𝐿=16 sequence. This performance is also obtained for a much longer preamble with 𝑁pr=1024 repetitions and slightly improved with 𝑁pr=4096 at the cost of increased preamble energy (cf. Figure 5), much longer signals (see Table 1), and much higher processing effort. From these results, it seems inefficient for EDs to choose extremely long preambles with 𝑁pr1024.

The IEEE 802.15.4a standard also defines different channels with specific bandwidths and carrier frequencies. An analysis of the various channels is given in the next section.

5.2. Effect of Frequency Channels

The IEEE 802.15.4a standard defines 16 channels in three frequency bands, the subgigahertz band (<1 GHz), the low band (3.2–4.8 GHz), and the high band (5.9–10.3 GHz). The channel bandwidths 𝐵 range from 499.2 to 1354.97 MHz. As mentioned in Section 3, keep in mind that the CEPT allows only the usage of the frequency bands 3.1–4.8 and 6–8.5 GHz for UWB, where, for LDC, the signals have to be shorter than 5 ms. Thus, the short preamble symbol with spreading 𝐿=64 and 𝑁pr=4096 is not allowed for LDC transmission. It is well known that a higher carrier frequency 𝑓𝑐 causes higher losses and thus less received signal strength according to Friis’ equation. A larger bandwidth leads to a higher allowed transmit power (see Section 3). To evaluate this tradeoff, six channels are analyzed in this paper, using the short preamble codes with 𝐿=64.

Figure 7 shows the relation between input and output SNR for the specific channels. As seen from (9), the CR is again independent of the channel bandwidths. For the ED, variations occur due to the different pulse and receiver bandwidths. It can be observed that the channels with the large bandwidths need up to 1.5dB more 𝐸𝑝/𝑁0 in the working point to achieve the same LSNR.

Figure 7: Relation between input SNR 𝐸LOS/𝑁0 and output SNR LSNR for the short preamble codes with a spreading 𝐿=64 and 𝑁pr=16 with respect to the IEEE 802.15.4a channels. The channel parameters in the legend are in GHz.

Figure 8 shows the allowed preamble energies for the specific channels. The larger bandwidths of the preambles allow a gain of up to 4dB, which is sufficient to compensate the SNR loss of the ED shown in Figure 7. This is also seen from (10), where the equivalent bandwidth 𝑊RRC influences LSNR linearly and the additional energy improves the SNR quadratically in the working point. Thus, a gain of up to 2.5dB can be achieved. For the CR, the additional energy will directly improve the performance.

Figure 8: Allowed preamble energy for the specific IEEE802.15.4a channels.

Figure 9 shows the maximal operating distance and the maximal allowed pathloss for the specific channels. It can be observed that the low-band channels (Ch 3 and Ch 4) perform better than the high-band channels due to lower free-space losses. Only half the operating range is obtained when 𝑓𝑐 is increased from 4 to 8 GHz. A shift in the carrier frequency leads to a change of 𝐸LOS/𝑁0, but it does not change the relation of input and output SNR (compare Ch 3 and Ch 9). A shift of the bandwidth changes this relation due to a change of ND, which is observable for Ch 9 and Ch 11. The operating range is doubled with the CR when the bandwidth is increased from 500 MHz to 1.33 GHz, while only the 1.3 fold distance is achieved with the ED. The best performance is obtained at Ch 4, which has a low carrier frequency 𝑓𝑐4GHz and a large bandwidth of 𝐵1.33GHz. It reaches a 𝑑max10620m (PLmax88dB) for the CR and 𝑑max620m for the ED (PLmax63dB). The mandatory Ch 3 of the low band shows a significantly better performance in comparison to the mandatory channel in the high band Ch 9 due to the lower 𝑓𝑐.

Figure 9: Maximal operating distance and maximal allowed pathloss for the specific IEEE 802.15.4a channels. The dashed lines show the characteristic for a channel with 𝐵=499.2MHz with the ND of Figure 6.

6. Conclusions

A coherent receiver and an energy detector have been studied for ranging in IEEE 802.15.4a, in the sense of maximal allowed transmit energy, maximal operating distance, and maximum allowed pathloss.

The maximal allowed transmit energy according to the FCC/CEPT regulations depends strongly on the parameters of the preamble. For most of the preamble code sequences, the average power limit applies. A longer spreading of the preamble symbols leads to a performance gain, because a larger preamble energy is obtained. As the FCC/CEPT limits the power spectral density, a higher bandwidth leads to an increased energy too.

The maximal operating distance is calculated from the link budget. The coherent receiver directly depends on the receiver input SNR, while the energy detector is also influenced strongly by the parameters of the preamble codes due to the noncoherent combining loss. A 64-symbol repetitions preamble is most efficient for the energy detector due to lower noncoherent combining losses and the short preamble symbols are preferable due to less despreading effort. The channels from the low-frequency band achieve longer ranges due to the lower pathloss. The mandatory low-frequency channel (𝑓𝑐=4.5GHz) achieves almost twice the range in comparison to the mandatory high-frequency channel (𝑓𝑐=8GHz). A gain is obtained for the high-bandwidth channels. The range is almost doubled with the coherent receiver, while the energy detector reaches only a gain of 30 percent.

The low-complexity energy detector achieves maximal operating distances of several hundred meters, while the coherent receiver reaches distances up to several thousand meters in free-space. Thus, both receiver architecture are appropriate for real-time locating systems and sensor networks in typical indoor scenarios.


The authors would like to warmly thank G. Kubin, TU Graz, and M. Pistauer, CISC Semiconductor GmbH, for their support in the project. The project was funded by the Federal Ministry for Transport, Innovation and Technology (BMVIT) and the Austrian Research Promotion Agency (FFG).


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