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VLSI Design

Volume 2008 (2008), Article ID 738983, 6 pages

http://dx.doi.org/10.1155/2008/738983

## Wave Pipelining Using Self Reset Logic

Department of Electrical Engineering, School of Engineering, Santa Clara University, 500 El Camino Real, Santa Clara, CA 95053, USA

Received 1 May 2007; Accepted 9 December 2007

Academic Editor: Jean-Baptiste Begueret

Copyright © 2008 Miguel E. Litvin and Samiha Mourad. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

#### Abstract

This study presents a novel design approach combining wave pipelining and self reset logic, which provides an elegant solution at high-speed data throughput with significant savings in power and area as compared with other dynamic CMOS logic implementations. To overcome some limitations in SRL art, we employ a new SRL family, namely, dual-rail self reset logic with input disable (DRSRL-ID). These gates depict fairly constant timing parameters, specially the width of the output pulse, for varying fan-out and logic depth, helping accommodate process, supply voltage, and temperature variations (PVT). These properties simplify the implementation of wave pipelined circuits. General timing analysis is provided and compared with previous implementations. Results of circuit implementation are presented together with conclusions and future work.

#### 1. Introduction

Wave pipelining (WP) is a suitable solution for fast arithmetic circuit implementation since it renders high throughput, while reducing the area and power overhead in a pipeline by removing intermediate registers. Such registers result in a latency penalty due to their setup and clock-to-output times and introduce delays for each stage. The area savings are realized due to (a) area devoted to the registers themselves, and (b) area needed for the clock distribution and buffering to control such registers.

In WP designs,
each stage holds its output just enough time to guarantee that the next stage
will be able to capture the data and start the computation of its own outputs
to the following element in the pipe. So ideally, data have to progress *simultaneously* through the stages to achieve the maximum data throughput. Specific timing
constraints apply to guarantee no data corruption. These designs are
essentially asynchronous, but can be synchronized by the use of input and
output registers (implemented with latches, or flip-flops) as long as timing
conditions are met so that outputs are captured at an appropriate time. This
circuit arrangement is shown in Figure 1.

Self reset logic (SRL) provides circuit implementations where “everything is quiet” when no new data are received. For single-rail implementations, power consumption is “data dependent.” In a dual-rail implementation, there will be pulses propagating along the wave path (either at the direct or inverse outputs of each stage); every time new data is presented at primary inputs. We introduce a new family of dual-rail SRL with input disable (DRSRL-ID). A typical cell is shown in Figure 2. In Section 2, we discuss its operation. Section 3 describes the timing constraints that apply when designing wave-pipelining circuits with DRSRL-ID. Section 4 presents an application circuit, and Section 5 presents conclusions and future work.

#### 2. Self Reset Logic

A DRSRL-ID buffer-inverter cell is shown in Figure 2. The gate will generate an output pulse
at the direct output (buf) or the inverse output (inv) only if the inputs validate the logic function *F* or its inverse; otherwise, both outputs will remain at zero. Once inputs evaluation starts, the gate disconnects the inputs for the duration of the cycle time *τ* defined as follows: where the following definitions hold:*denotes data delay forward* (the
time from the leading edge of the input data transition that validates *F*
or *FN* to the leading edge of the pulse at the output);*denotes width* of the *output pulse*;*denotes recovery
time* (the time
elapsed from the trailing edge of the output pulse to the trailing edge of the
reset pulse). For inputs to be
evaluated, they have to be active for a minimum overlapping time, ,
that must be longer than the capture time .

* denotes* capture time, the time from
the leading edge of the input data transition that validates *F* or *FN*
to the falling edge of the pulse at the internal summing node (*SUM or *).

We prefer to refer to
the output pulse (at either output) so the condition is written as The width of the output pulse depends strongly on the characteristics of the output stage of the gate, but is independent of the loading while *fan-out* is equal to or less than 8 (for the gate families we have worked with). When a set of inputs validates the logic function *F* (or *Fn*), the corresponding output pulse starts only after the delay ,
but its width depends on the delay through the feedback loop, which postcharges the summing nodes. Also, since we disable inputs, once an output pulse starts, is also *independent of the width of input pulses,* while they satisfy condition (2). Recovery time and delay forward can also be made equal for a family of gates. Then, the *cycle time*
will be a constant for the circuit implemented with these gates. It defines the minimum clock period at which new data can be pushed into the combinational circuit when received from an input register.

Figure 4 shows these timing parameters depicted for idealized waveforms corresponding to the behavior of a typical gate of this type. The outputs *Y* and correspond to outputs buf and inv
of the buffer-inverter cell shown in Figure 2. For a complete description and characterization of these gates, we refer the reader to [1, 2].

An XOR/XNOR gate is shown in Figure 3. In this case, the use of shared elements
between the FN and FNb blocks minimizes the number of devices needed. This
approach is especially useful when implementing more complex gates.
Additionally, in this case, we show the implementation of the self reset
without using the extra inverters in the feedback loop. In the present case, as
we actually use the internal reset *pulse_rst* signal to disable
input readout as well as to control the postcharge of the summing nodes, we can
safely play with the width of the resetting pulse, without being affected by
the switching activity at the gate inputs.

#### 3. Wave Pipelining with DRSRL-ID

The wave-pipelining circuit is an
asynchronous structure, which can be made to work in a synchronous structure by
adding an input and an output register, controlled by clock, as shown in Figure 1. This requires careful selection of
the timing parameters. In the rest of this section, we explain the relationship
between these different timing parameters using the following symbols:number of data waves in the pipeline;global clock;period of the global clock;clock at input register;clock at output register;total latency (time elapsed from launching a data wave from the input register until the corresponding result arrives at the output register); maximum delay through the combinational logic;maximum path delay difference through the combinational logic;phase shift between and ;phase shift between and ;: : constructive skew (phase shift between the clocks that control the launching and receiving registers);register clock-to-*Q* delay;register setup time;register hold time;uncontrollable clock skew. The width of the output pulse of the input register must satisfy the following: The timing conditions are where is the worst timing difference expected at any given stage. Then, considering also the delay forward , that is,
As can be observed in Figure 5, the total latency is Analyzing the situation corresponding to a “late arriving” pulse versus an “early arriving” one, as shown in Figure 6, one can demonstrate that for wave pipelining with DRSRL-ID,

Comparing with regular WP CMOS implementation, as shown by [3–5], in that case the conditions for safe pipelining include the following (11) and (12): Condition (12) is a two-sided constraint on , , and , showing the behavior as we sweep frequencies: Condition (12) applies to the regular WP circuits and defines a set of “valid intervals” of frequencies at which the circuit will behave in a wave-pipelining mode. The higher the frequency, the narrower the valid clocking interval [4, 5]. So it becomes extremely important to control delays carefully, and there is a strong dependency on the process, voltage, and temperature (PVT) conditions.

In contrast, in DRSRL-ID there is a maximum frequency at which the circuit can operate in WP mode, as stated by conditions (6) and (7). As long as the clock period is greater than the cycle time *τ* of the gates, early arrival data from data wave (*N* + 1) will not interfere with late arrivals from the previous data wave. Trying to operate at higher frequencies will generate a situation where, at a given stage, bits would arrive “too early” and will be ignored by that stage, since these gates will still have their inputs disabled. For DRSRL-ID, at frequencies below what condition (7) states, the combinational logic will still function properly (with different *k* values). The only difficulty resides in capturing the computation result at the output register. Such behavior can be obtained by adding a latch at the end of the combinational logic, which will update when and if new data arrives, that is, converting the last stage into a static one. These characteristics render the technique presented in this paper very desirable.

The conditions on in DRSRL-ID are similar to the conditions on for CMOS WP, rendering a theoretic lower data rate. In other words, we could design for a suboptimal frequency, but building headroom for process, voltage, and temperature (PVT), that is, we accept a maximum operating frequency, and design with a built-in margin.

Observe that condition (4) implies that we need to minimize the timing difference among signals arriving at any given stage , since this directly impacts the maximum achievable operating frequency. One still needs to do “rough tuning” to equalize timing paths at each stage: add gates to shorter paths, and maintain a solid layout engineering that looks into equalizing wire loads. The “fine-tuning” proposed in other implementations [4] may not add much in this case because of the “built-in” headroom by the gates. (Fine-tuning refers to the careful resizing of gates at transistor level, according to the needs of each signal path.)

The method described in this paper renders a stable circuit that may meet all specifications on the first approach, at the price of having added this extra margin in the gates themselves.

#### 4. An Illustrative Example

##### 4.1. Wave-Pipelining Parallel Multiplier

Amultiplier was used to illustrate the concepts. It was implemented in a 1.2 V–0.18 *μ*m CMOS process using a library of DRSRL-ID cells. The multiplier consists of three major blocks: the partial product generator (), the partial product reducer (), and an adder (*carry propagate*). Figure 7 shows a block diagram.

In the first stage, the partial products () are generated. Each is the product of each bit of the multiplier by every bit of the multiplicand. Thus, for an multiplication, *n* (*n*-bit wide) are generated. These have to be added to obtain the final result.

The next stage is the partial product reducer (), which reduces the *n*
of an *n*-bit multiplier to two, hence the name of *reducer*. This is the main block of the multiplier, which we have implemented as a Wallace tree using *carry-save adders* ().Timing of the cell has been adjusted so that the delay forward of both outputs (S, Co) is approximately the same. The two final elements are added by means of an adder to generate the final result. We have used the *carry look-ahead* structure proposed in [5], with a slight modification to control the fan-out and the loading at critical points [2].

This block by itself is essentially asynchronous. We have added input and output registers for timing analysis when the multiplier is inserted in a synchronized pipeline. Theregisters were implemented by a set of edge-triggered flip-flops. The output register must sample the final stage of the adder while the result pulses are available. For this wave-pipelining application, all paths have been equalized using “rough padding,” that is, adding buffers to the shorter paths to get the same number of stages in all cases.

To make the delays through the longest and shortest paths through the logic as close as possible, we tried to tightly control the difference in arrival times of all signals connected to a certain stage. This implies not only the delay equalization of the different elements of logic at a given stage, but also the delay equalization of the interconnects between successive stages and controlling the total loading of a given intermediate driver. A careful layout plan is important, but in the present design there is a certain tolerance level for differences in arrival times. This is true as long as one can guarantee that all valid input pulses at a given stage will overlap long enough to generate the output pulse within the time frame of valid inputs for the next stage.

##### 4.2. Simulation Results and Analysis

Results of spice simulation of
8-bit multiplier, implemented in a 0.18 *μ*m
CMOS process, running at 2.5 GHz data rate, are shown in Figure 8. It can be
observed that as the pulse waves advance through the stages of the multiplier,
the timing difference among signals at a given stage is minimal, so they conform to a coherent
data wave. Here, the following signals are depicted: the global ideal clock *clk*,
the output , together with inputs and . Since the inputs shown leave a clock cycle in
between, where all input bits are made zero, for clarity, it is easy to observe
two nonzero input patterns, before the first output is shown. The pattern shown
corresponds to decimal products: (), (), (), (), and (),
alternated with () for power analysis.

In Figure 8, the global clock signal is shown as reference; it is the almost perfect rectangular wave. At the end of its first pulse, we see the overlapping input signals, entering the multiplier. As mentioned above, in this case, inputs remain at zero, during the next clock cycle, and so we do not observe any other activity. At the next clock cycle, the 2nd set of nonzero inputs is applied, but only within the following pulse of the clock we see the first set of outputs (very close together, and almost completely enclosed within that clock pulse).

A new set of nonzero inputs will enter the pipelined circuit, before the multiplier outputs the result corresponding to the second nonzero set of inputs. Actually, the multiplier is able to calculate a new result at each clock tick; we have just interleaved a set of zero-valued inputs in between for clarity.

The maximum
timing difference among output bits occurs in this design between bits (early arrival) and (late arrival) and is approximately 74
picoseconds = . The maximum delay through
the combinational logic is 978 picoseconds. The delay through
the FF (timing difference between and *ckira* is 52 picoseconds = .

Here, = 400 picoseconds, that is, = 2.5 GHz. Setup time = 20 picoseconds, and hold time = 50 picoseconds.

Looking at signals between different stages in the compressor, we have measured the following:(i) = 210 picoseconds, = 89 picoseconds,(ii); = 101 picoseconds, at the slowest path, and(iii): = (210 + 89 + 101) picoseconds= 400 picoseconds.

Since the input register is always sending pulses, by means of the direct outputs Q or the inverse ones Qn, then the power consumption is average no matter what pattern is presented at the inputs. Whenever single rail implementations are possible, there will be power savings, since pulses will be generated at gate outputs only if input signals validate the gate logic function, but gate outputs will remain at zero otherwise.

It is worth noting that as the width of the multiplier grows, the total latency increases, but the data throughput remains unchanged, as far as we can control the wire loading, since the maximum operating frequency depends on the cycle time of the gates.

#### 5. Conclusions and Proposed Future Work

Wave pipelining is especially suitable for designs that show a high degree of parallelism and regularity. If that were not the case, the circuit has to be first transformed to achieve such parallelism.The design shown provides a practical proof of the feasibility of using the proposed technique in many applications, where pipelining is suitable. Wave pipelining provides savings in area and timing, since all intermediate storage elements are removed from the circuit, saving also from the point of view of timing overhead. The use of self reset logic provides savings in power and area with respect to a comparable CMOS dynamic implementation, since clock distribution for dynamic gates is avoided, as shown in [2], where a comparison was made between two implementations of an adder: domino logic versus DRSRL-ID. The use of dual-rail self reset logic with input disable functionality (DRSRL-ID) has additional advantages, providing a fairly constant pulse width, and in so doing avoiding “pulse-width adjusting structures” [6]. It provides an additional tolerance in the design for differences in arrival times of signals at any stage. While such tolerance is built-in in the structure of the gate family, it comes at the price of adding to the total cycle time and affects the minimum clock period used to pump in new data into the circuit. The reduction in area and power savings, plus the simplified equalization mechanism due to the built-in tolerance, makes this approach suitable for many fast processing designs.

Additionally, if we use as the last stage an
SR-latch, which will only be updated each time new data has arrived, then, we
are making the last stage “static,” and in so doing, we can reduce the
operating frequency as we need to interface with the next stage (moving the
design from a *k*-wave mode to a single
wave, if so needed). At the same time, we must maintain constraint (3) on the
width of the input pulse to the first stage implemented with DRSRL-ID. The
recommended approach would be to use a pulse generator, which will generate one
pulse at the valid input clock transition.

The basic DRSRL-ID is suitable for structures with feedback, and this is an area we will investigate further.There is also special interest in asynchronous circuit applications. The DRSRL-ID application shown here uses the simplest protocol: “just sending data” and sacrifices elasticity for higher throughput. Many variations are possible according to circuit needs.

#### Acknowledgment

M. E. Litvin is an IEEE regular member and S. Mourad is an IEEE Fellow. They thank Dr. Fabian Klass for his invaluable comments and advise during the research period.

#### References

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