Single Magnetic Element-Based High Step-Up Converter for Energy Storage and Photovoltaic System with Reduced Device Count

Arif, Muhammad; Shahzad, Mohsin; Li, Qianmu; Saleem, Jawad; Aslam, Muhammad Shamrooz; Majid, Abdul

doi:https://doi.org/10.1155/2020/5892190

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Abstract Introduction Results and Discussion Conclusions Data Availability Conflicts of Interest Acknowledgments References Copyright Related Articles

Research Article | Open Access

Volume 2020 | Article ID 5892190 | https://doi.org/10.1155/2020/5892190

Single Magnetic Element-Based High Step-Up Converter for Energy Storage and Photovoltaic System with Reduced Device Count

Muhammad Arif,¹Mohsin Shahzad,¹Qianmu Li,^2,3,4Jawad Saleem,¹Muhammad Shamrooz Aslam,⁵and Abdul Majid¹

Academic Editor: Marcio Eisencraft

Received26 May 2020

Accepted29 Sept 2020

Published24 Oct 2020

Abstract

The multiport DC-DC power converter is a prominent area of research in power electronics due to its highly dense design, reduced device count, and high energy efficiency. In this paper, a nonisolated single magnetic element-based high step-up three-port converter for an energy storage system is presented. The proposed converter has two input ports and one output port. The coupled inductor with switched capacitor is used to achieve high voltage gain. The key features of the proposed converter are high conversion gain, low voltage stress, zero voltage switching (ZVS), and zero current switching (ZCS). The detailed theoretical analysis and operation of the converter are elaborated. The energy efficiency of the proposed converter is calculated and compared with the other counterparts. Ansys Maxwell is used for the coupled inductor finite element modeling. To verify the applicability and functionality of proposed converter, a converter with two inputs ( and ) and one output at is tested in the laboratory.

1. Introduction

The photovoltaic (PV) power generation, owing to its high availability and cleanliness, is rapidly increasing with a total global capacity of at the end of year 2019 [1]. The PV generation system can be operated in islanded and grid-connected mode. The islanded PV system is an optimal choice for the remote area electricity generation and distribution. However, the unpredictability of the solar energy and the electric load demand pose a challenge for the broad level exploitation of solar energy. Therefore, the PV framework needs a battery to remunerate these challenges. On the contrary, to produce AC, the inverter requires input voltage of 380– on its DC link. As the voltage generated by the PV system is less than , a step-up converter is required to keep DC link voltage constant. To address the aforementioned challenges, the power electronic converters provide best solution and efficient interface to PV source [2]. Conventionally, a separate unidirectional converter for PV is used along with a battery connected to either side of converter using bidirectional DC-DC converter. This configuration of separate converters for each source increases the system volume and cost at lower system efficiency. Therefore, various multiport converter topologies are reported in literature [3]. A multiport converter is a converter with central controls, inputs/outputs, and bidirectional ports [4]. Using multiport converter instead of standard single converters builds the system proficiency, improves components execution, and diminishes the volume and cost [2–4].

A special type of multiport converter is the three-port converter (TPC) including a unidirectional port for power source, a bidirectional port for battery system, and the output port. Recent advances and topologies used for the integration of renewable sources are discussed in [5]. There are some challenges that need to be addressed in TPC such as minimum device count, high voltage gain, optimal control for power sharing, and smooth transition between operating modes. As an optimal design of the TPC, it is always desired that the converter should have minimum device count and high conversion gain. In traditional step-up converters, duty cycle is increased to achieve large step-up ratio. Increasing duty cycle beyond certain limit results in increased switching loss. The coupled inductors, transformers, multipliers, and switched capacitors are some of the various techniques used to achieve high voltage gain [5, 6].

Three-port converter topologies are classified as fully isolated [7–11], partly isolated where one of the ports is isolated [12–18], and nonisolated [19–33]. The isolated structures are suitable for the high power and low frequency applications, but efficiency of these structures is usually low. The partly isolated structures are good with prominent features of high power density and efficiency. The nonisolated three-port converter topologies are preferred because of reduced size, cost, and high power density [5]. The nonisolated topologies are more suitable for standalone PV and low voltage distribution and storage system [3–5]. Recent advancements in nonisolated structures are presented in [19–33]. In [19], a series-parallel connection based on the H-bridge nonisolated converter has been proposed. A converter having input of and output of at a frequency of is discussed. The converter topology can be modified by increasing its gain and reducing the device count. A nonisolated TPC converter using two inductors and switched capacitors and diodes has been proposed in [20]. The converter has comparatively high conversion gain and low voltage stress on the main operating switch. However, the converter has too many components and inductive elements leading to reduced system efficiency. A three-port high gain converter for the HEV’s applications has been addressed in [22]. To accomplish high gain, a coupled inductor along with series of the capacitors has been used. However, the topology can further be optimized by reducing the number of device count. In [23], a high gain converter using a Cockcroft–Walton multiplier method has been proposed. The converter has bigger size due to many passive components. In [24], a coupled inductor-based high step-up TPC with high conversion gain for PV integration has been proposed. However, the use of two coupled inductors results in increased size and cost. A soft switching high gain TPC with the merits of soft switching and demerits of high component count and two coupled inductors is discussed in [25]. A very simple topology of TPC based on the coupled inductor for PV system is given in [26]. The topology used only coupled inductor, and its gain depends on the turn ratio of the coupled inductor. In [27–29], the authors presented modeling and control of a three-port converter for various applications.

To attain high conversion gain, one coupled inductor is used in nonisolated converter topology [30, 31]. A coupled inductor-based high gain converter is better than isolation transformer due to simpler winding structures and lower conduction losses [6]. In [30], a three-port high gain converter using a coupled inductor is introduced. The converter has the merit of high conversion gain, reduced voltage stress on the main operating switch, and soft switching. However, the converter has a coupled inductor and single inductor which increases its size. A coupled inductor with series capacitor-based high gain TPC with comparatively reduced device count and only one magnetic element has been proposed in [31]. This topology can be further improved by minimizing the device count and better design of the magnetic element. Most of the topologies reported in the literature are switched at less than . The switching frequency of theses converter topologies can be increased and control schemes can be improved.

In this paper, a three-port bidirectional converter using one coupled inductor and switched capacitor is proposed. The switching frequency is increased up to in order to shrink the size, increase power compactness, and improve the dynamic response of the converter. The proposed converter has high conversion gain and soft switching capability for the main operating switches. The analysis of the converter is performed in continuous conduction mode (CCM). The operating principle and corresponding theoretical analyses of the proposed converter in different modes are discussed in detail. The presented work is based on [32, 33], with the following vital modifications and extensions:(1)Novel converter topology with reduced device count(2)Reduction in size and cost by increasing switching frequency up to (3)Finite element modeling of coupled inductor in order to improve the design(4)Operation of converter in different modes(5)Loss and efficiency analyses(6)Development of lab prototype and presentation of measured results

The introduction and comprehensive summary of various converter topologies are presented in Section 1, followed by the operation and analysis of the proposed converter in Section 2. The detailed proposed design and the control scheme are elaborated in Section 3 and 4, respectively. The experimental results, loss and efficiency analyses, and comparison are given in Section 5. Lastly, the overall work is concluded in Section 6.

2. Operation and Analysis of the Proposed Topology

The proposed converter, comprised of one unidirectional port for PV and two bidirectional ports, is presented in Figure 1. The proposed converter comprises four active switches , , , and and four diodes , , , and . The proposed converter has three capacitors: clamp capacitor , switch capacitor , and output capacitor . The proposed converter structure has only one coupled inductor. The coupled inductor along with switch capacitor is used to achieve high voltage conversion gain at output port. In order to diminish the voltage stress on the fundamental switch , the active clamp circuit is used. The primary winding of the coupled inductor is common between the inputs, i.e., battery and PV system. The proposed converter has the bidirectional power flow ability along with intersource power-sharing. The switch is the main operating switch and remains active most of the time. The switches and are used for boost and buck operations, respectively. The switch is used to control power between the PV and battery system. The converter operation is discussed for single-input single-output (SISO), double-input single-output (DISO), and single-input double-output (SIDO) modes. Large value capacitors are used to maintain constant voltage. The detailed operation of the converter in different operating modes is explained in the following sections.

2.1. Converter Operation in SISO Mode

In SISO mode, there are two modes of operation; buck and boost. In boost operation, the battery/PV provides power to the load, whereas in buck operation, the battery is charged from the DC link capacitor.

2.1.1. Boost Operation

In this mode, the power is transferred to the load either by the battery or from the PV. For a single switching cycle, the converter has five operating states. The theoretical switching waveform and current flow paths in CCM are presented in Figure 2(a). The detailed operation of the converter in five states is illustrated.

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State-I [] (Figure 3(a)): this is a very short interval; the switch is turned on at , is on, and the switches and are off. The current increases linearly and it is equal to the magnetizing inductance current at time . The net current passing through the primary side is . The secondary side current is equal to which decreases linearly and is equal to zero at time . The diodes , , and are reverse biased. State-II [] (Figure 3(b)): in this interval, both the switches and are on. The current is increasing linearly. The current is also increasing linearly but in opposite direction. As the current direction and voltage polarity are reversed on both sides of the coupled inductor, the capacitor is charging the switched capacitor through the diode . The current increases until it equals at time . State-III [] (Figure 3(c)): at time , the switch is turned off. The leakage inductance energy is transferred to the capacitor . The voltage over the switch is clamped to . The currents and through the diodes and decrease. As this is a very short interval, the current direction and voltage polarities over the coupled inductor remain unchanged. State-IV [] (Figure 3(d)): for this interval, the switch is on and the switch is off. The current direction and the voltage polarities change over the coupled inductor. The diode is forward biased and current decreases. At time , the current become zero, and the diode is reverse biased. State-V [] (Figure 3(e)): the switch is off, whereas switch is still on. At time , the diode is turned off. The current direction is the same in both primary and secondary windings of the coupled inductor. The magnetizing current is decreasing. The energy from the battery, magnetizing inductor , and the leakage inductor along with the capacitor is delivered to the load.

2.1.2. Buck Operation

In SISO mode, the input is main DC link capacitor and the output is battery. The large value DC link capacitor is used. In this mode, the main operating switch is . The theoretical waveforms for this mode are presented in Figure 2(b). For one complete cycle, the operation the converter is explained for the following time intervals. State-I []: this is a very short interval; the switch is turned on. The magnetizing current and leakage current are decreasing and the switch is turned on at almost zero current flowing through it. State-II [] (Figure 4(a)): in this interval, the switch is on. The coupled inductor magnetizing current is increasing in opposite direction. The current is charging the inductor . The interval ends at time . State-III [] (Figure 4(b)): at the end of time , the switch is turned off. Initially, current increases abruptly and then becomes zero at time . This interval ends at . State-IV [] (Figure 4(c)): at time , the diode conducts and switch is off. The current increases and becomes zero at time . State-V [] (Figure 4(d)): in this interval, the switch is still off. The magnetizing current charges the battery through the body diode . This interval ends at time .

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2.2. Converter Operation in DISO Mode

During this mode, the PV power is not enough to fulfill the load requirement. The additional power is supplied to the load from the battery by controlling the switch . During this dual-input single-output (DISO) mode, both PV and battery provide power to load. The theoretical waveforms and current flow paths are presented in Figures 2(c) and 5, respectively. The operation of the converter during this mode is explained in six states. State-I [] (Figure 5(a)): at time , switches and are turned on. The leakage current is almost zero, and the current is changing its direction. The switches , and are turned on under almost zero current. This state is similar to state-1 of the SISO mode. All the relationships and equations are same for this state. State-II [] (Figure 5(b)): this state is similar to state-II of the SISO mode. All the relationships and equations are same for this state. State-III [] (Figure 5(c)): the switch is still on while at the end of time , the switch is turned off. The diode is forward biased, and the current increases. The current decreases linearly and is zero at time . The primary voltage of the coupled inductor has been decreased from its previous value. State-IV [] (Figure 5(d)): in this state, inductor is charged through by the PV source. The switch is still on and the current increases linearly with a slope of . State-V [] (Figure 5(e)): at the end of time , the switch is turned off and the diode is forward biased. The voltage over the switch clamps to the voltage . State-VI [] (Figure 5(f)): all the switches are off, PV is delivering power to the load, and voltage is boosted to . This state is similar to state-V of the SISO mode except the switch is off and diode is turned on.

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2.3. Converter Operation in SIDO Mode

In this mode, the PV generation is greater than the load requirement and excess energy is used to charge the battery. The theoretical switching characteristics are presented in Figure 2(d). The switches and are active in this mode. The operation of the converter in different states is discussed as follows. State-I [] (Figure 6(a)): at time , the switches and are turned on. The current decreases linearly and tends to reach zero, consequently turning on both and at ZCS. The operation of the converter in this state is similar to its operation in state-1 of SISO mode. State-II [] (Figure 6(b)): in this state, the operation of the converter is similar to the operation of state-1 of SISO mode except the source. All relationships and equations are valid for this state. State-III [] (Figure 6(c)): at time , the switch is turned off while the switch is still on. The current and the magnetizing current decrease linearly. At time , the current becomes zero. State-IV [] (Figure 6(d)): at time , the switch is still on. The leakage and magnetizing currents are equal while the secondary current is zero. The magnetizing current is used to charge the battery through the diode and switch . The battery charge current is controlled by the duty cycle . The slope of is equal to . The state ends at time when the switch is turned off. State-V [] (Figure 6(e)): the switches and are turned off. The diode conducts, transferring the leakage inductance energy to the clamping capacitor . This state ends at time . State-VI [] (Figure 6(f)): all switches are off. The operation of the converter in this state is similar to state-V of the SISO mode. The only difference is that the diode conducts and the source is PV. All the relationships and equations are similar to state-V of SISO mode for this state.

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3. Steady-State Analysis and Design Considerations

Averaging of the converter in each mode is performed by applying voltage-second balance on magnetizing current equations, whereas the ampere-second method is used on all capacitors’ voltage equations. The steady-state analysis results for each mode are elaborated in SISO, DISO, and SIDO modes.

3.1. SISO Mode

3.1.1. Boost Operation

By applying the voltage-second balance on the magnetizing inductance , the voltage over the magnetizing inductor is calculated according to the following equation:

The voltages across the capacitors and are computed according to equations (2) and (3), respectively.

The steady-state voltage across the capacitor is computed by using the following equation:

The output voltage across the load is computed by using the following equation:

The converter gain increases significantly by increasing the duty cycle and the turn ratio “n” of the coupled inductor. Change in the output voltage with the change in duty cycle in SISO mode is presented in Figure 7(a). The comparison of output voltage and the input voltage at various values of the turn ratio “n” is presented in Figure 7(b). This shows that the output voltage increases linearly with the input voltage. The proposed converter has the gain of 8.

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From the analysis, it is observed that for both the inputs (PV and battery), the voltage across the load will remain the same. The switch is used to control the output voltage for both the inputs PV and battery.

3.1.2. Buck Operation

In this mode, the battery is charged from the main DC link capacitor. The main operating switch is S3. For the steady-state operation in this mode, the voltage-second balance on the magnetizing current is applied. The steady-state voltage, when the battery is charged from the DC link capacitor, is given in the following equation:where .

3.2. DISO Mode

For this mode, the voltage-second balance on the magnetizing inductance is applied. The voltage across the magnetizing inductance is computed by using the following equation:

The voltage across the capacitor is computed by the following equation:

The voltage across the capacitor is determined by the following equation:where .

The output voltage across the capacitor is computed by using the following equation:where .

3.3. SIDO Mode

For the calculation of conversion ratio, we are applying voltage-second balance. In this mode, the only input is PV. To calculate the conversion ratio, voltage-second balance is applied on the magnetizing inductance . The voltage across is calculated by using the following equation:

The voltage across the capacitors , , and is calculated by using equations (12)–(14), respectively.

The output voltage over the capacitor is calculated by using the following equation:

3.4. Design Consideration

The design specifications of the proposed converter are given in Table 1. The main components used in the converter design are coupled inductor, power MOSFETs, capacitors, and diodes. The appropriate selection of coupled inductor, capacitors, and power semiconductor devices is very crucial for the desired operation of the converter.

3.4.1. Coupled Inductor Design

The required parameters of the coupled inductor used in the converter topology are given in Table 1. The equivalent inductance is , where is the mutual inductance. The number of turns for the required inductance value of the is calculated by using equation (15). The required peak inductor current is , and the duty cycle . For , the core ETD-39 and wire AWG-14 are selected. The peak primary inductor current is calculated using equation (16). The peak to peak primary inductor current is . The peak current through diode is calculated by using equation (17). The relationship between and is expressed in equation (18). The required values of the coupled inductor are and for DCM operation. For the CCM operation, inductor values are and .where N67 material is used which has the following properties: , , , and = maximum flux density.

The primary inductance value and its losses are calculated by using equations (20) and (21), respectively. The magnetizing inductance can be calculated by using equation (22).

The secondary inductor value is calculated by using the relationship , where . Here, is the number of turns on the primary side and is the number of turns on the secondary side. The coupling coefficient between the primary and secondary inductors is calculated by using equation (23), where as the power loss in secondary inductor is calculated by equation (24).

The leakage inductance value on the primary side can be calculated by equation (25), where is the resonance frequency and the is the output capacitance of the switch .

The finite element model of the coupled inductor is developed in Ansys Maxwell 3D by using PExprt tool. The coupled inductor’s Ansys analysis report for both the DCM and CCM operations is given in Tables 2 and 3, respectively. In Table 2, the coupled inductor performance results are shown for the DCM mode. The results show the primary and secondary inductance ( and ) values of and , respectively, with turn ratio . The current density is found to be for and for with the losses of and for and , respectively. The value of is found to be and for and , respectively. The total losses for DCM mode are , and window filling is and for and , respectively. Similarly, the coupled inductor performance results for CCM mode are shown in Table 3. The values of the current density, coupled inductor, losses in primary and secondary windings are shown. The DC resistance value of the used wire is = . Moreover, the losses in the core and winding are expressed as and , respectively. Furthermore, the window occupancy for both primary and secondary winding is obtained as and , respectively.

The finite element model (FEM) of coupled inductor for magnetic flux density is presented in Figure 8(a), whereas the FEM for the magnetic flux lines is presented in Figure 8(b).

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3.4.2. Capacitor Selection

Three capacitors used in this topology are the clamping capacitor , the switched capacitor , and DC link capacitor . Equation (26) gives the relationship between the clamp capacitor and the input voltage .

The blocking voltage over the switch is expressed in the following equation:

The minimum value of the capacitors and is calculated by using equations (28) and (29), where as the value of the output capacitor is determined by equation (30). The voltage across the capacitor is determined by equation (31).

3.4.3. Selection of Switches and Diodes

The switches , , and have low voltage blocking capability while switch has high voltage blocking capability. The maximum voltage stress across the main switch and the diode is calculated by the following equation:where is the input voltage at PV or battery port. The switch and the diode bear maximum voltage stress, and it can be expressed as follows:

The voltage stress across the switch and the diode is expressed as follows:whereas the voltage stress across the switch and the diode is given as follows:

The maximum current stress across the switches and diodes of the proposed converter occurs if only one source is serving the load. The current stress on the switches and and the diodes and is equal to as expressed in equation (18). The current stress on the switch is equal to and is express in equation (20). The peak current through the diode is = , where is expressed in equation (17).

A fast recovery diode is selected for , and an ultrafast recovery diode is chosen for , whereas the ordinary rectifier diodes with required current and voltage blocking capability are selected for and . The ratings of components used in simulation or prototype are given in Table 1.

4. Control Scheme and Operational Mode Selection

Generally, output of the converter is regulated to satisfy the load requirement and also for the constant input to the inverter. The objective of the control scheme is to regulate the main DC link capacitor voltage in SISO (boost operation) and DISO modes. In SIDO mode, the voltage over the main DC link capacitor is regulated at constant value. The charging and discharging battery is also regulated in this mode. In SISO (buck operation), the objective is to direct the battery voltage and control the charging current . The charging and discharging battery is also regulated in this mode. The mode selection and pulse width modulation (PWM) block is presented in Figure 9(a). The flow diagram explains the power flow and conditions for the transition in different modes. In this figure, , , , and are the gate driving/control signals for switches , , , and , respectively. The control signal is the main control signal applied to switch in all modes. There are separate control loops for the input and output voltage ports. The comp-1 regulator keeps the PV system power at the maximum value while comp-2 regulates the output voltage at . The activity mode is resolved by present working conditions, for example, load power, battery condition of charge, and accessible PV power. The control algorithm which determines the operational mode is presented in Figure 9(b). Initially, values of all terminal variables, i.e., , , , and , are acquired. If the battery is fully charged and PV power is not available, then the converter will operate in SISO (boost) mode. The converter operates in SISO (buck) mode when the PV power is not available and there is light load.

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If both the PV power and load are available, the converter will operate in DISO Mode. As the converter is developed for the standalone PV system, the equation is always true, and hence the intermittence in PV power is always compensated by the battery system. Moreover, the converter is operated in SIDO mode; otherwise, the battery charge protection is active. The switch is used to control the battery power. The transition between modes is very smooth. For example, in SIDO mode, if with constant load, is used to control the battery power until . The converter operates in SISO mode (buck) and battery is charged from the main DC link capacitor , and the control signal is . This ensures smooth and soft transition between the ports.

5. Results and Discussion

In order to examine the exhibition of proposed topology and hypothetical investigation, a converter is tested in the laboratory. The proposed converter has high gain in SISO, DISO, and SIDO modes. The complete analytical and experimental results for SISO, DISO, and SIDO modes are presented. Main parts of the converter are common to all the ports. The parameters of different parts utilized in this model are given in Table 1. There are only minor changes in the values of parameters; however, shapes of the current and voltage waveforms remain the same. Design and selection of the components is performed by considering the possible maximum rating values. As the converter has only one coupled inductor, can be expanded by keeping the duty cycle constant in order to increase the gain of the converter.

5.1. Experimental Results

A hardware prototype is developed and tested to prove the operational concepts of the proposed converter. The photographs of laboratory workstation and converter topology are presented in Figures 10(a) and 10(b), respectively. A four-layer PCB is developed by using Altium Designer, with one power plan and one segmented ground plan. In the design of PCB, the number of controlled interfaces is reduced. This reduces the common mode voltage between the interface ports so that there is less coupling from the cables into or out of the system. In order to minimize the return current path impedance, the return current path is kept closer to the signal path. “Moats” in PCB ground plan are avoided. All power and ground rails are carefully checked to ensure they do not offer common impedance routes within or outside the unit. The parameters and particulars of different segments utilized in equipment model are expressed in Table 1. For the generation of control signals, TI Launchpad-F28379D is used. The instruments used for measurements are GDS-810C, a measuring module (USM-3IV), an oscilloscope, and a multimeter.

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The measured waveforms of the converter for SISO boost operating modes are presented in Figure 11. The magnetizing current and gate driving signal () are presented in Figure 11(a). The duty cycle for the switch is 0.6. The peak to peak voltage is . The gate driving signal () vs coupled inductor leakage current is presented in Figure 11(b), whereas the gate driving signal () and coupled inductor secondary current is presented in Figure 11(c).

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Because of the leakage inductance of coupled inductor, the switch is operated under ZCS. The leakage inductance is calculated according to equation (26). The measured series resonance inductance which is combination of leakage inductance and stray inductance is . The equivalent capacitance obtained from the output capacitance of primary and synchronous switches is . The switch is also operated under almost ZVS condition without using the external inductors and capacitors. There are some ripples in the measurement of the current due to the low sensitivity of the measuring module. The measured gate signal of switch and currents are presented in Figure 11(d), whereas the vs curves are plotted in Figure 11(e) which are in close similarity with the theoretical results plotted in Figure 2(a).

The gate signal of switch S₁ and corresponding drain source voltage are plotted in Figure 12. It is obvious from the results that drain source voltage of S₁ becomes zero before application of gate signal. In this case, almost ZVS is achieved by the series resonance tank L_lk-C_outs1 tank circuit. The tank circuit consists of the primary side leakage inductance of the coupled inductor and the output capacitance C_outs1 of the switch S₁. During the off time of the signal , the L_lk-C_outs1 circuit resonates. As the output capacitance of the switch S₁ has been discharged by the series resonant circuit, this results in power savings and improvement in the energy efficiency of the converter.

The measured results for the SISO mode (buck operation) are presented in Figure 13. The duty cycle is equal to 0.3. The gate signal and magnetizing current are presented in Figure 13(a). There are some ripples in the current due to the measuring setup and coupled inductor leakage inductance. The gate signal of switch and corresponding drain source voltages are presented in Figure 13(b). It is obvious from the figure that switch is operated under almost ZVS. In Figure 13, switching gate signal and inductor’s secondary current are presented. During the turn on of the switch , both currents and are same.

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The measured results of converter in DISO mode are presented in Figure 14. The duty cycle of switch is 0.6, and the duty cycle of switch is 0.5. The gate signals and and the current are presented in Figure 14(a). Battery supplies power to the load as long as the switch is on. As the battery voltage drops below the threshold voltage, the switch is turned off and power is supplied to the load by the source . Due to this, all of the results are same as those of the SISO (boost operation) mode. In Figure 14(b), the gate signals and and magnetizing current are presented. The secondary current and gate signals and are presented in Figure 14(c). It is observed that the only voltage rating and polarity of and affect the performance of the converter.

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The measured results of converter in SIDO mode are plotted in Figure 15. The duty cycle of switch is 0.6, and the duty cycle of the switch is 0.68. The control signals and and the magnetizing current are presented in Figure 15(a). The switch is used to control the battery current . The excess energy generated by PV is used to charge the battery. The control signals and and drain source voltage are presented in Figure 15(b). The gate signals and and secondary current are expressed in Figure 15(c). The diode is used to transfer the voltage stress of the switch . The gate signals and and the current are presented in Figure 15(d). The current control signals and are presented in Figure 15(e). The control signals and and current are presented in Figure. 15(f)

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5.2. Loss and Efficiency Analysis

The loss and efficiency of the proposed converter are analyzed in each operating mode. The SISO boost mode of operation is the main operating mode. For this mode, the efficiency of the converter as a function of output power is depicted in Figure 16(a). The converter has maximum efficiency of at the power of with in SISO mode. For input , the converter has maximum efficiency of 95.5%. The converter efficiency is calculated by using equation (36). Table 4 shows the measured values of the input (rms) current, output current, resistive load, and efficiency values. The converter efficiency increases with the output power up to . Beyond this point, the energy efficiency remains constant. Under these conditions, the losses in the converter are estimated and illustrated in the pie diagram in Figure 16(b). At the output power of , the efficiency of the converter is with input voltage and % with the input voltage . The maximum contribution in the loss is due to the main operating switch , i.e., FQA34N20 with . The switch loss can be calculated by using equation (37). The coupled inductor has the second highest losses with the exact value of . The series capacitor contributes about 15% and input diode has the share of 10% in the total losses. Rest of the components have relatively low loss contribution.

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For the DISO mode of operation, the main power loss is the same as that of the SISO mode, but there is additional power loss due to the diode and the switch . The losses due to diode and switch are calculated by using equation (38) and equation (39), respectively.where is the discharging current of the battery.

The power loss in the SIDO mode is investigated by considering the loss of the diodes and and the switch . The conduction losses for the diodes and are computed by using equation (38) and equation (40), respectively. The switch conduction loss can be computed by using equation (41).

5.3. Comparison Study

The proposed converter is compared with several similar converters suitable for standalone PV system in Table 5. The comparison of proposed converter with [24, 30, 31] is performed. The proposed converter has higher switching frequency and uses less number of components to construct more useful features such as bidirectional power flow ports.

The proposed converter has only one conversion stage with common coupled inductor for both inputs. It has only one magnetic element, i.e., coupled inductor, four MOSFETs, two capacitors, and two diodes, whereas the converter in [24] has two coupled inductors, two inductors, and five switches. Similarly, the converter in [30] has one coupled inductor and one inductor along with three MOSFETs, six diodes, and three capacitors. The size of the converter is relatively small. The converter proposed in [31] has one coupled inductor, three MOSFETs and capacitors each, and six diodes. The converter has no extra inductor.

The conversion gain in SISO mode is presented in Figure 16(c). The gain of the proposed converter is almost equal to the converter explained in [31], but with minimum number of device count and size. In terms of device count, operational modes, and efficiency, the proposed converter outperforms its competitors. The operational comparison of the converters in terms of the efficiency in different operating modes is presented in Figure 16(d). In comparison with other counterparts by considering the input voltage 48V, the proposed converter has the maximum efficiency of which is the same as that of [24, 31], but the proposed converter is operated at while others are operated at .

A topological comparison of the proposed converter along with its other recent counterparts is presented in Table 6. The advantages and disadvantages of the various recent structures are given. In [8], the converter provides good galvanic isolation, ZVS operation, and high conversion gain. The converter has large size, and this makes the converter costly with lower efficiency. The converter is good for the AC/DC integrated DC microgrid applications. In [10], the proposed converter has single conversion stage for each port which makes the control scheme simpler. It also has the ZVS and ZCS operation for all the switches. However, the large size and low efficiency limit the converter applications. In [12], the converter has the merits of high conversion gain and flexible operation between different modes. However, the converter has large size leading to the low efficiency. Merits and demerits of some other converters [13–15, 18] are summarized in Table 6.

The coupled inductor-based topologies are discussed in [24, 30, 31]. Brief overview of these topologies is given in Table 6, whereas detailed comparative analysis is given in Table 5. These topologies are preferred for the renewable energy integration with DC microgrid.

The proposed converter has the merits of high gain, simple structure, low voltage stress on the main switches, and high efficiency. The only challenging part is the need of decoupling control as most of the components are common in the structure which can be overcome with suitable control scheme.

6. Conclusions

A three-port bidirectional power converter using a coupled inductor and a switched capacitor is proposed. Size of the converter is reduced by increasing the switching frequency. Higher voltage gain is achieved by using active clamp and switch capacitor technique. Operation of the converter is illustrated in three distinct modes, i.e., SISO, DISO, and SIDO. The converter has high gain in all operational modes. The use of the clamp circuit results in the reduction of voltage stress on the switches. All the main operating switches are operated under ZVS and ZCS. The converter efficiency is calculated and loss analysis is performed using analytic, simulated, and experimental models. The proposed converter has efficiency in SISO mode. A prototype of a 100 W converter is developed by using and inputs and output voltage. The measured results of the converter are in close comparison with simulation results.

Data Availability

The data used to support the findings of this study are available from the corresponding author upon request.

Conflicts of Interest

The authors declare that they have no conflicts of interest.

Acknowledgments

This study was supported in part by the 2019 Industrial Internet Innovation and Development Project from Ministry of Industry and Information Technology of China, the 2020 Industrial Internet Innovation and Development Project from Ministry of Industry and Information Technology of China, the 2018 Jiangsu Province Major Technical Research Project “Information Security Simulation System,” and the Fundamental Research Funds for the Central Universities (30918012204). This study was also supported by starting PhD fund of Guangxi University of Science and Technology (no. 20Z14).

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Copyright © 2020 Muhammad Arif et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

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